US20080095377A1 - Btsc encoder - Google Patents
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- US20080095377A1 US20080095377A1 US11/927,742 US92774207A US2008095377A1 US 20080095377 A1 US20080095377 A1 US 20080095377A1 US 92774207 A US92774207 A US 92774207A US 2008095377 A1 US2008095377 A1 US 2008095377A1
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04H—BROADCAST COMMUNICATION
- H04H20/00—Arrangements for broadcast or for distribution combined with broadcast
- H04H20/86—Arrangements characterised by the broadcast information itself
- H04H20/88—Stereophonic broadcast systems
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- G—PHYSICS
- G10—MUSICAL INSTRUMENTS; ACOUSTICS
- G10L—SPEECH ANALYSIS OR SYNTHESIS; SPEECH RECOGNITION; SPEECH OR VOICE PROCESSING; SPEECH OR AUDIO CODING OR DECODING
- G10L19/00—Speech or audio signals analysis-synthesis techniques for redundancy reduction, e.g. in vocoders; Coding or decoding of speech or audio signals, using source filter models or psychoacoustic analysis
- G10L19/008—Multichannel audio signal coding or decoding using interchannel correlation to reduce redundancy, e.g. joint-stereo, intensity-coding or matrixing
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04N—PICTORIAL COMMUNICATION, e.g. TELEVISION
- H04N5/00—Details of television systems
- H04N5/44—Receiver circuitry for the reception of television signals according to analogue transmission standards
- H04N5/60—Receiver circuitry for the reception of television signals according to analogue transmission standards for the sound signals
- H04N5/602—Receiver circuitry for the reception of television signals according to analogue transmission standards for the sound signals for digital sound signals
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04N—PICTORIAL COMMUNICATION, e.g. TELEVISION
- H04N7/00—Television systems
- H04N7/06—Systems for the simultaneous transmission of one television signal, i.e. both picture and sound, by more than one carrier
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04R—LOUDSPEAKERS, MICROPHONES, GRAMOPHONE PICK-UPS OR LIKE ACOUSTIC ELECTROMECHANICAL TRANSDUCERS; DEAF-AID SETS; PUBLIC ADDRESS SYSTEMS
- H04R5/00—Stereophonic arrangements
- H04R5/04—Circuit arrangements, e.g. for selective connection of amplifier inputs/outputs to loudspeakers, for loudspeaker detection, or for adaptation of settings to personal preferences or hearing impairments
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04S—STEREOPHONIC SYSTEMS
- H04S5/00—Pseudo-stereo systems, e.g. in which additional channel signals are derived from monophonic signals by means of phase shifting, time delay or reverberation
- H04S5/02—Pseudo-stereo systems, e.g. in which additional channel signals are derived from monophonic signals by means of phase shifting, time delay or reverberation of the pseudo four-channel type, e.g. in which rear channel signals are derived from two-channel stereo signals
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Abstract
Description
- The patent application is a divisional of co-pending U.S. application Ser. No. 09/638,245, filed Aug. 14, 2000, which is a continuation of U.S. application Ser. No. 09/041,244, filed Mar. 12, 1998, now U.S. Pat. No. 6,118,879, issued Sep. 12, 2000, which is a divisional of U.S. application Ser. No. 08/661,412, filed Jun. 7, 1996, now U.S. Pat. No. 5,796,842, issued Aug. 18, 1998. The contents of each of the earlier applications are hereby incorporated by reference as recited herein in their entirety.
- The present invention relates generally to stereophonic audio encoders used for television broadcasting. More particularly, the invention relates to a digital encoder for generating the audio signals used in the broadcast of stereophonic television signals in the United States and in other countries.
- In the 1980's, the United States Federal Communications Commission (FCC) adopted new regulations covering the audio portion of television signals which permitted television programs to be broadcast and received with bichannel audio, e.g., stereophonic sound. In those regulations, the FCC recognized and gave special protection to a method of broadcasting additional audio channels endorsed by the Electronic Industries Association and the National Association of Broadcasters and called the Broadcast Television Systems Committee (BTSC) system. This well known standard is sometimes referred to as Multichannel Television Sound (MTS) and is described in the FCC document entitled, MULTICHANNEL TELEVISION SOUND TRANSMISSION AND AUDIO PROCESSING REQUIREMENTS FOR THE BTSC SYSTEM (OET Bulletin No. 60, Revision A, February 1986), as well as in the document published by the Electronic Industries Association entitled, MULTICHANNEL, TELEVISION SOUND BTSC SYSTEM RECOMMENDED PRACTICES (EIA Television Systems Bulletin No. 5, July 1985). Television signals generated according to the BTSC standard are referred to hereinafter as “BTSC signals”.
- The original monophonic television signals carried only a single channel of audio. Due to the configuration of the monophonic television signal and the need to maintain compatibility with existing television sets, the stereophonic information was necessarily located in a higher frequency region of the BTSC signal making the stereophonic channel much noisier than the monophonic audio channel. This resulted in an inherently higher noise floor for the stereo signal than for the monophonic signal. The BTSC standard overcame this problem by defining an encoding system that provided additional signal processing for the stereophonic audio signal. Prior to broadcast of a BTSC signal by a television station, the audio portion of a television program is encoded in the manner prescribed by the BTSC standard, and upon reception of a BTSC signal a receiver (e.g., a television set) then decodes the audio portion in a complementary manner. This complementary encoding and decoding insures that the signal-to-noise ratio of the entire stereo audio signal is maintained at acceptable levels.
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FIG. 1 is a block diagram of a prior art BTSC encoding system, or more simply, aBTSC encoder 100, as defined by the BTSC standard.Encoder 100 receives left and right channel audio input signals (indicated inFIG. 1 as “L” and “R”, respectively) and generates therefrom a conditioned sum signal and an encoded difference signal. It should be appreciated that while the system of the prior art and that of the present invention is described as useful for encoding the left and right audio signals of a stereophonic signal that is subsequently transmitted as a television signal, the BTSC system also provides means to encode a separate audio signal, e.g., audio information in a different language, which is separated and selected by the end receiver. Further, noise reduction components of the BTSC encoding system can be used for other purposes besides television broadcast, such as for improving audio recordings. -
System 100 includes aninput section 110, a sumchannel processing section 120, and a differencechannel processing section 130.Input section 110 receives the left and right channel audio input signals and generates therefrom a sum signal (indicated inFIG. 1 as “L+R”) and a difference signal (indicated inFIG. 1 as “L−R”). It is well known that for stereophonic signals, the sum signal L+R may be used by itself to provide monophonic audio reproduction and it is this signal that is decoded by existing monophonic audio television sets to reproduce sound. In stereophonic sets, the sum and difference signals can be added to and subtracted from one another to recover the original two stereophonic signals (L) and (R).Input section 110 includes twosignal adders adder 114 subtracts the right channel audio input signal from the left channel audio input signal to generate the difference signal. As described above, the sum signal L+R is transmitted through a transmission media with the same signal to noise ratio as achieved with the prior monophonic signals. However, the difference signal L−R is transmitted though a very noisy channel, particularly at the higher frequency portion of the relevant spectrum so that the decoded difference signal has a poorer signal-to-noise ratio because of the noisy medium and reduced dynamic range of the medium. The dynamic range is defined as the range of signals between the level of the noise floor and the maximum level where signal saturation occurs. In the difference signal channel the dynamic range decreases at higher frequencies. Accordingly, the difference signal is subjected to additional processing than that of the sum signal so that the dynamic range can be substantially preserved. - More particularly, the sum
channel processing section 120 receives the sum signal and generates therefrom the conditioned sum signal.Section 120 includes a 75 μspreemphasis filter 122 and abandlimiter 124. The sum signal is applied to the input offilter 122 which generates therefrom an output signal that is applied to the input ofbandlimiter 124. The output signal generated by the latter is then the conditioned sum signal. - The difference
channel processing section 130 receives the difference signal and generates therefrom the encoded difference signal.Section 130 includes a fixed preemphasis filter 132 (shown implemented as a cascade of twofilters variable gain amplifier 134 preferably in the form of a voltage-controlled amplifier, a variable preemphasis/deemphasis filter (referred to hereinafter as a “variable emphasis filter”) 136, an overmodulation protector andbandlimiter 138, afixed gain amplifier 140, abandpass filter 142, anRMS level detector 144, afixed gain amplifier 146, abandpass filter 148, anRMS level detector 150, and areciprocal generator 152. - The difference signal is applied to the input of
fixed preemphasis filter 132 which generates therefrom an output signal that is applied vialine 132 d to an input terminal ofamplifier 134. An output signal generated byreciprocal generator 152 is applied vialine 152 a to a gain control terminal ofamplifier 134.Amplifier 134 generates an output signal by amplifying the signal online 132 d using a gain that is proportional to the value of the signal online 152 a. The output signal generated byamplifier 134 is applied via line 134 a to an input terminal ofvariable emphasis filter 136, and an output signal generated byRMS detector 144 is applied vialine 144 a to a control terminal offilter 136.Variable emphasis filter 136 generates an output signal by preemphasizing or deemphasizing the high frequency portions of the signal on line 134 a under the control of the signal online 144 a. The output signal generated byfilter 136 is applied to the input of overmodulation protector andbandlimiter 138 which generates therefrom the encoded difference signal. - The encoded difference signal is applied via
feedback path 138 a to the inputs offixed gain amplifiers amplifier 140 is applied to an input ofbandpass filter 142 which generates therefrom an output signal that is applied to the input ofRMS level detector 144. The latter generates an output signal as a function of the RMS value of the input signal level received fromfilter 142. The amplified signal generated byamplifier 146 is applied to the input ofbandpass filter 148 which generates therefrom an output signal that is applied to the input ofRMS level detector 150. The latter generates an output signal as a function of the RMS value of the input signal level received fromfilter 148. The output signal ofdetector 150 is applied vialine 150 a toreciprocal generator 152, which generates a signal online 152 a that is representative of the reciprocal of the value of the signal online 150 a. As stated above, the output signals generated byRMS level detector 144 andreciprocal generator 152 are applied tofilter 136 andamplifier 134, respectively. - As shown in
FIG. 1 , the differencechannel processing section 130 is considerably more complex than the sumchannel processing section 120. The additional processing provided by the differencechannel processing section 130, in combination with complementary processing provided by a decoder (not shown) receiving a BTSC signal, maintains the signal-to-noise ratio of the difference channel at acceptable levels even in the presence of the higher noise floor associated with the transmission and reception of the difference channel. Differencechannel processing section 130 essentially generates the encoded difference signal by dm compressing, or reducing the dynamic range of the difference signal so that the encoded signal may be transmitted through the limited dynamic range transmission path associated with a BTSC signal, and so that a decoder receiving the encoded signal may recover all the dynamic range in the original difference signal by expanding the compressed difference signal in a complementary fashion. The differencechannel processing section 130 is a particular form of the adaptive signal weighing system described in U.S. Pat. No. 4,539,526, which is known to be advantageous for transmitting a signal having a relatively large dynamic range through a transmission path having a relatively narrow, frequency dependent, dynamic range. - Briefly, the difference channel processing section may be thought of as including a wide
band compression unit 180 and aspectral compression unit 190. The wideband compression unit 180 includesvariable gain amplifier 134 preferably in the form of a voltage controlled amplifier, and the components of the feedback path for generating the control signal to amplifier 134 and comprisingamplifier 146,band pass filter 148,RMS level detector 150, andreciprocal generator 152.Band pass filter 148 has a relatively wide pass band, weighted towards lower audio frequencies, so in operation the output signal generated byfilter 148 and applied toRMS level detector 150 is substantially representative of the encoded difference signal.RMS level detector 150 therefore generates an output signal online 150 a representative of a weighted average of the energy level of the encoded difference signal, andreciprocal generator 152 generates a signal online 152 a representative of the reciprocal of this weighted average. The signal online 152 a controls the gain ofamplifier 134, and since this gain is inversely proportional to a weighted average (i.e., weighted towards lower audio frequencies) of the energy level of the encoded difference signal, wideband compression unit 180 “compresses”, or reduces the dynamic range, of the signal online 132 a by amplifying signals having relatively low amplitudes and attenuating signals having relatively large amplitudes. - The
spectral compression unit 190 includesvariable emphasis filter 136 and the components of the feedback path generating a control signal to thefilter 136 and comprisingamplifier 140,band pass filter 142 andRMS level detector 144. Unlikefilter 148,band pass filter 142 has a relatively narrow pass band that is weighted towards higher audio frequencies. As is well known, the transmission medium associated with the difference portion of the BTSC transmission system has a frequency dependent dynamic range and the pass band offilter 142 is chosen to correspond to the spectral portion of that transmission path having the narrowest dynamic range (i.e., the higher frequency portion). In operation the output signal generated byfilter 142 and applied toRMS level detector 144 contains primarily the high frequency portions of the encoded difference signal.RMS level detector 144 therefore generates an output signal online 144 a representative of the energy level in the high frequency portions of the encoded difference signal. This signal then controls the preemphasis/deemphasis applied byvariable emphasis filter 136 so in effect thespectral compression unit 190 dynamically compresses high frequency portions of the signal on line 134 a by an amount determined by the energy level in the high frequency portions of the encoded difference signal as determined by thefilter 142. The use of thespectral compression unit 190 thus provides additional signal compression towards the higher frequency portions of the difference signal, which combines with the wideband compression provided by thevariable gain amplifier 134 to effectively cause more overall compression to take place at high frequencies relative to the compression at lower frequencies. This is done because the difference signal tends to be noisier in the higher frequency part of the spectrum. When the encoded difference signal is decoded with a wideband expander and a spectral expander in a decoder (not shown), respectively in a complementary manner to the wideband compression unit 180 andspectral compression unit 190 of the encoder, the signal-to-noise ratio of the L−R signal applied to the differencechannel processing section 130 will be substantially preserved. - The BTSC standard rigorously defines the desired operation of the 75
μs preemphasis filter 122, the fixedpreemphasis filter 132, thevariable emphasis filter 136, and thebandpass filters amplifiers amplifier 134,RMS level detectors reciprocal generator 152. The BTSC standard also provides suggested guidelines for the operation of overmodulation protector andbandlimiter 138 andbandlimiter 124. Specifically,bandlimiter 124 and the bandlimiter portion of overmodulation protector andbandlimiter 138 are described as low pass filters with cutoff frequencies of 15 kHz, and the overmodulation protection portion of overmodulation protector andbandlimiter 138 is described as a threshold device that limits the amplitude of the encoded difference signal to 100% of full modulation where full modulation is the maximum permissible deviation level for modulating the audio subcarrier in a television signal. - Since
encoder 100 is defined in terms of mathematical descriptions of idealized filters it may be thought of as an idealized or theoretical encoder, and those skilled in the art will appreciate that it is virtually impossible to construct a physical realization of a BTSC encoder that exactly matches the performance oftheoretical encoder 100. Therefore, it is expected that the performance of all BTSC encoders will deviate somewhat from the theoretical ideal, and the BTSC standard defines maximum limits on the acceptable amounts of deviation. For example, the BTSC standard states that a BTSC encoder must provide at least 30 db of separation from 100 Hz to 8,000 Hz where separation is a measure of how much a signal applied to only one of the left or right channel's inputs appears erroneously in the other of the left or right channel's outputs. - The BTSC standard also defines a composite stereophonic baseband signal (referred to hereinafter as the “composite signal”) that is used to generate the audio portion of a BTSC signal. The composite signal is generated using the conditioned sum signal, the encoded difference signal, and a tone signal, commonly referred to as the “pilot tone” or simply as the “pilot”, which is a sine wave at a frequency fH where fH is equal to 15,734 Hz. The presence of the pilot in a received television signal indicates to the receiver that the television signal is a BTSC signal rather than a monophonic or other non BTSC signal. The composite signal is generated by multiplying the encoded difference signal by a waveform that oscillates at twice the pilot frequency according to the cosine function cos(4πfHt), where t is time, to generate an amplitude modulated, double-sideband, suppressed carrier signal and by then adding to this signal the conditioned sum signal and the pilot tone.
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FIG. 2 is a graph of the spectrum of the composite signal. InFIG. 2 the spectral band of interest containing the content of the conditioned sum signal (or the “sum channel signal”) is indicated as “L+R”, the two spectral sidebands containing the content of the frequency shifted encoded difference signal (or the “difference channel signal”) are each indicated as “L−R”, and the pilot tone is indicated by the arrow at frequency fH. As shown inFIG. 2 , in the composite signal the encoded difference signal is used at 100% of full modulation, the conditioned sum signal is used at 50% of full modulation, and the pilot tone is used at 10% of full modulation. - Stereophonic television has been widely successful, and existing encoders have performed admirably, however, virtually every BTSC encoder now in use has been built using analog circuitry technology. These analog BTSC encoders, and particularly the analog difference channel processing sections, due to their increased complexity have been relatively difficult and expensive to construct. Due to the variability of analog components, complex component selection and extensive calibration have been required to produce acceptable analog difference channel processing sections. Further, the tendency of analog components to drift, over time, away from their calibrated operating points has also made it difficult to produce an analog difference channel processing section that consistently and repeatably performs within a given tolerance. A digital difference channel processing section, if one could be built, would not suffer from these problems of component selection, calibration, and performance drift, and could potentially provide increased performance.
- Further, the analog nature of existing BTSC encoders has made them inconvenient to use with newly developed, increasingly popular, digital equipment. For example, television programs can now be stored using digital storage media such as a hard disk or digital tape, rather than the traditional analog storage media, and in the future increasing use will be made of digital storage media. Generating a BTSC signal from a digitally stored program now requires converting the digital audio signals to analog signals and then applying the analog signals to an analog BTSC encoder. A digital BTSC encoder, if one could be built, could accept the digital audio signals directly and could therefore be more easily integrated with other digital equipment.
- While a digital BTSC encoder would potentially offer several advantages, there is no simple way to construct an encoder using digital technology that is functionally equivalent to the
idealized encoder 100 defined by the BTSC standard. One problem is that the BTSC standard defines all the critical components ofidealized encoder 100 in terms of analog filter transfer functions. As is well known, while it is generally possible to design a digital filter so that either the magnitude or the phase response of the digital filter matches that of an analog filter, it is extremely difficult to match both the amplitude and phase responses without requiring large amounts of processing capacity for processing data sampled at very high sampling rates or without significantly increasing the complexity of the digital filter. Without increasing either the sampling frequency or the filter order, the amplitude response of a digital filter can normally only be made to more closely match that of an analog filter at the expense of increasing the disparity between the phase responses of the two filters, and vice versa. However, since small errors in either amplitude or phase decrease the amount of separation provided by BTSC encoders, it would be essential for a digital BTSC encoder to closely match both the amplitude and phase responses of an idealized encoder of the type shown at 100 inFIG. 1 . - For a digital BTSC encoder to provide acceptable performance, it is critical to preserve the characteristics of the analog filters of an
idealized encoder 100. Various techniques exist for designing a digital filter to match the performance of an analog filter; however, in general, none of these techniques produce a digital filter (of the same order as the analog filter) having amplitude and phase responses that exactly match the corresponding responses of the analog filter.Ideal encoder 100 is defined in terms of analog transfer functions specified in the frequency domain, or the s-plane, and to design a digital BTSC encoder, these transfer functions must be transformed to the z-plane. Such a transformation may be performed as a “many-to-one” mapping from the s-plane to the z-plane which attempts to preserve time domain characteristics. However, in such a transformation the frequency domain responses are subject to aliasing and may be altered significantly. Alternatively, the transformation may be performed as a “one-to-one” mapping from the s-plane to the z-plane that compresses the entire s-plane into the unit circle of the z-plane. However, such a compression suffers from the familiar “frequency warping” between the analog and digital frequencies. Prewarping can be employed to compensate for this frequency warping effect, however, prewarping does not completely eliminate the deviations from the desired frequency response. These problems would have to be overcome to produce a digital BTSC encoder that performs well and is not unduly complex or expensive. - There is therefore a need for overcoming the difficulties and developing a digital BTSC encoder.
- It is an object of the present invention to substantially reduce or overcome the above-identified problems of the prior art.
- Another object of the present invention is to provide an adaptive digital weighing system.
- Still another object of the present invention is to provide an adaptive digital weighing system for encoding an electrical information signal of a predetermined bandwidth so that the information signal can be recorded on or transmitted through a dynamically-limited, frequency dependent channel having a narrower dynamically-limited portion in a fast spectral region than in at least one other spectral region of the predetermined bandwidth.
- And another object of the present invention is to provide a digital BTSC encoder.
- Yet another object of the present invention is to provide a digital BTSC encoder that prevents ticking, a problem that can arise with substantially zero input signal levels.
- And another object of the present invention is to provide a digital BTSC encoder that uses a sampling frequency that is a multiple of a pilot tone signal frequency of 15,734 Hz so as to prevent interference between the signal information of the encoded signal with the pilot tone signal.
- Still another object of the invention is to provide a digital BTSC encoder for generating a conditioned sum signal and an encoded difference signal that include substantially no signal energy at the pilot tone frequency of 15,734 Hz.
- Yet another object of the present invention is to provide a digital BTSC encoder including a sum channel processing section for generating the conditioned sum signal, and a difference processing section for generating the encoded difference signal, the sum channel processing section including devices for introducing compensatory phase errors into the conditioned sum signal to compensate for any phase errors introduced into the encoded difference signal by the difference channel processing section.
- And another object of the present invention is to provide a digital BTSC encoder including a digital variable emphasis unit, the unit including a digital variable emphasis filter characterized by a variable coefficient transfer function, and the unit further including a device for selecting the coefficients of the variable coefficient transfer function as a function of the signal energy of the encoded difference signal.
- Yet another object of the present invention is to provide a digital BTSC encoder including a composite modulator for generating a composite modulated signal from the conditioned sum signal and the encoded difference signal.
- Still another object of the present invention is to provide a digital BTSC encoder that may be implemented on a single integrated circuit.
- These and other objects are provided by an improved BTSC encoder that includes an input section, a sum channel processing section, and a difference channel processing section all of which are implemented using digital technology. In one aspect, the input section includes high pass filters for preventing the BTSC encoder from exhibiting “ticking”. In another aspect, the BTSC encoder uses a sampling frequency that is equal to an integer multiple of the pilot frequency.
- In yet another aspect, the sum channel processing section generates a conditioned sum signal, and the difference channel processing section generates an encoded difference signal, and the sum channel processing section includes components for introducing a phase error into the conditioned sum signal to compensate for any phase errors introduced into the encoded difference signal by the difference channel processing section.
- According to yet another aspect, the invention provides an adaptive digital weighing system for encoding an electrical information signal of a predetermined bandwidth so that the information signal can be recorded on or transmitted through a dynamically-limited, frequency dependent channel having a narrower dynamically-limited portion in a fast spectral region than in at least one other spectral region of the predetermined bandwidth.
- Still other objects and advantages of the present invention will become readily apparent to those skilled in the art from the following detailed description wherein several embodiments are shown and described, simply by way of illustration of the best mode of the invention. As will be realized, the invention is capable of other and different embodiments, and its several details are capable of modifications in various respects, all without departing from the invention. Accordingly, the drawings and description are to be regarded as illustrative in nature, and not in a restrictive or limiting sense, with the scope of the application being indicated in the claims.
- For a fuller understanding of the nature and objects of the present invention, reference should be had to the following detailed description taken in connection with the accompanying drawings in which the same reference numerals are used to indicate the same or similar parts wherein:
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FIG. 1 shows a block diagram of a prior art idealized BTSC encoder; -
FIG. 2 shows a graph of the spectrum of the composite signal generated in accordance with the BTSC standards; -
FIG. 3 shows a block diagram of one embodiment of a digital BTSC encoder constructed according to the invention; - FIGS. 4A-C show block diagrams of low pass filters used in the digital BTSC encoder shown in
FIG. 3 ; -
FIG. 5 shows a detailed block diagram of the wideband compression unit used in the digital BTSC encoder shown inFIG. 3 ; -
FIG. 6 shows a block diagram of the spectral compression unit used in the digital BTSC encoder shown inFIG. 3 ; -
FIG. 7 shows a flow chart used for calculating the filter coefficients of the variable emphasis filter used in the spectral compression unit shown inFIG. 6 ; - FIGS. 8A-D show block diagrams that illustrate signal scaling that may be used to preserve resolution and decrease the chance of saturation in fixed point implementations of digital BTSC encoders constructed according to the invention;
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FIG. 9 shows a detailed block diagram of the composite modulator shown in FIGS. 8B-C; and -
FIG. 10 shows a block diagram of one preferred embodiment of sum and difference channel processing sections that may be used in digital BTSC encoders constructed according to the invention. -
FIG. 3 is a block diagram of one embodiment of adigital BTSC encoder 200 constructed according to the invention.Digital encoder 200 is constructed to provide performance that is functionally equivalent to the performance of idealized encoder 100 (shown inFIG. 1 ). As withidealized encoder 100,digital encoder 200 receives the left and right channel audio input signals and generates therefrom the conditioned sum signal and the encoded difference signal, however, indigital encoder 200 these input and output signals are digitally sampled signals rather than continuous analog signals. - The choice of sampling frequency fs for the left and right channel audio input signals significantly affects the design of
digital encoder 200. In the preferred embodiments, the sampling frequency fs is chosen to be an integer multiple of the pilot frequency fH, so that fs=NfH where N is an integer, and in the most preferred embodiments, N is selected to be greater than or equal to three. It is important forencoder 200 to insure that the conditioned sum and encoded difference signals do not contain enough energy at the pilot frequency fH to interfere with the pilot tone that is included in the composite signal. As will be discussed in greater detail below, it is therefore desirable for at least some of the filters indigital encoder 200 to provide an exceptionally large degree of attenuation at′ the pilot frequency fH, and this choice of sampling frequency fs simplifies the design of such filters. -
Digital encoder 200 includes aninput section 210, a sumchannel processing section 220 and a differencechannel processing section 230. Rather than simply implementing the differencechannel processing section 230 using digital technology, all threesections digital encoder 200 respectively correspond to individual components inidealized encoder 100. In general, the components ofdigital encoder 200 have been selected so that their amplitude responses closely match the respective amplitude responses of their corresponding components inencoder 100. This often results in there being a relatively large difference between the phase responses of corresponding components. According to one aspect of the present invention, means are provided indigital encoder 200 for compensating for or nullifying these phase differences, or phase errors. As those skilled in the art will appreciate, relatively small phase errors in the differencechannel processing section 230 may be compensated for by introducing similar phase errors in the sumchannel processing section 220, and implementing the sum channel processing section using digital technology simplifies the introduction of such desired compensating phase errors. - The
input section 210 ofencoder 200 includes two high pass filters 212, 214, and twosignal adders high pass filter 212, the latter generating therefrom an output signal that is applied to positive input terminals ofadders high pass filter 214 which generates therefrom an output signal that is applied to a positive input terminal ofadder 216 and to a negative input terminal ofadder 218.Adder 216 generates a sum signal (indicated inFIG. 3 as “L+R”) by summing the output signals generated byfilters Adder 218 generates a difference signal (indicated inFIG. 3 as “L−R”) by subtracting the output signal generated byfilter 214 from the output signal generated byfilter 212.Input section 210 is therefore similar to input section 110 (shown inFIG. 1 ) however,section 210 additionally includes the two high pass filters 212, 214 and generates digital sum and difference signals. - High pass filters 212, 214 preferably have substantially identical responses and preferably remove D.C. components from the left and right channel audio input signals. As will be discussed in greater detail below, this D.C. removal prevents encoder 200 from exhibiting a behavior referred to as “ticking”. Since the audio-information content of the left and right channel audio input signals of interest is considered to be within a frequency band between 50 Hz and 15,000 Hz, removal of D.C. components does not interfere with the transmission of the information content of the audio signals.
Filters Filters - Referring again to
FIG. 3 , the sumchannel processing section 220 receives the sum signal and generates therefrom the conditioned sum signal. In particular, the sum signal is applied to a 75μs preemphasis filter 222. Thefilter 222 in turn generates an output signal that is applied to a staticphase equalization filter 228. Thefilter 228 generates an output signal that is applied to alow pass filter 224 ofsection 220 which in turn generates the conditioned sum signal. - The 75
μs preemphasis filter 222 provides signal processing that is partially analogous to the filter 122 (shown inFIG. 1 ) ofidealized encoder 100. The amplitude response offilter 222 is preferably selected to closely match that offilter 122. As will be discussed further below, means are preferably provided in differencechannel processing section 230 for compensation for any differences in the phase responses offilters filter 222 is implemented as a fast order IIR filter having a transfer function H(z) that is described by the formula shown in the following Equation (2). - Static
phase equalization filter 228 performs processing that is not directly analogous to any of the components in idealized encoder 100 (shown inFIG. 1 ). As will be discussed in greater detail below, staticphase equalization filter 228 is used to introduce phase errors that compensate for phase errors introduced bydifference processing section 230. Briefly, staticphase equalization filter 228 is preferably an “all-pass” filter having a relatively flat amplitude response and a selected phase response. In one preferred embodiment,filter 228 is implemented as a first order IIR filter having a transfer function H(z) that is described by the formula shown in the following Equation (3). -
Low pass filter 224 provides processing that is partially analogous to bandlimiter 124 (shown inFIG. 1 ) ofencoder 100.Low pass filter 224 preferably provides a flat amplitude response in a pass band of zero to 15 kHz and a relatively sharp cutoff above 15.Filter 224 also preferably provides an exceptionally large degree of attenuation at the frequency fH of the pilot tone (i.e., 15,734 Hz). By providing this exceptionally large degree of attenuation,filter 224 insures that the conditioned sum signal does not include enough energy at the pilot frequency fH to interfere with the pilot tone used in the composite signal. As discussed above, selecting the sampling frequency fs to be equal to an integer multiple of the pilot frequency fH simplifies the design of a filter that provides an exceptionally large degree of attenuation at the pilot frequency and therefore simplifies the design offilter 224.Filter 224 preferably has a null at the pilot frequency fH and preferably provides at least 70 dB of attenuation for all frequencies from the pilot frequency fH up to one-half the sample rate. -
FIG. 4A is a block diagram illustrating one preferred embodiment oflow pass filter 224. As shown inFIG. 4A , filter 224 may be implemented by cascading fivefilter sections filter sections
So in the embodiment shown inFIG. 4A ,filter 224 is tenth order IIR filter. - Referring again to
FIG. 3 , the differencechannel processing section 230 receives the difference signal and generates therefrom the encoded difference signal. The difference signal is applied to alow pass filter 238 a which generates therefrom an output signal that is applied to a fixedpreemphasis filter 232 a. The latter generates an output signal that is applied vialine 239 to an input terminal of awideband compression unit 280, and the encoded difference signal is applied viafeedback line 240 to a detector terminal ofwideband compression unit 280. The latter generates an output signal that is applied vialine 281 to an input terminal of aspectral compression unit 290, and the encoded difference signal is also applied viafeedback line 240 to a detector terminal ofunit 290. The latter generates an output signal that is applied to a fixedpreemphasis filter 232 b which in turn generates an output signal that is applied to aclipper 254.Clipper 254 generates an output signal that is applied to alow pass filter 238 b which in turn generates the encoded difference signal. - Low pass filters 238 a, 238 b, together form a low pass filter 238 that performs processing that is partially analogous to the bandlimiter portion of overmodulation protector and bandlimiter 138 (shown in
FIG. 1 ) ofidealized encoder 100. Preferably, filter 238 is implemented so that it is substantially identical tolow pass filter 224, which is used in the sumchannel processing section 220. Any phase errors introduced into the encoded difference signal by filter 238 are therefore compensated by balancing phase errors that are introduced into the conditioned sum signal byfilter 224. Filter 238 is preferably split into twosections - FIGS. 4B-C are block diagrams illustrating one preferred embodiment of the
respective filters FIG. 4B , filter 238 a may be implemented by cascading threefilter sections FIG. 4A ), and as shown inFIG. 4C , filter 238 b may be implemented by cascading twofilter sections filter 224. - Fixed preemphasis filters 232 a, 232 b (shown in
FIG. 3 ) together form a fixed preemphasis filter 232 that performs processing that is partially analogous to filter 132 (shown inFIG. 1 ) ofidealized encoder 100. The amplitude response of filter 232 is preferably selected to closely match the amplitude response offilter 132. In one embodiment, the phase responses offilters 232 and 132 are significantly different, and as will be discussed in greater detail below, the resulting phase errors are compensated for byfilters channel processing section 220. Filter 232 is preferably split into twosections - In one preferred embodiment, the difference between the phase responses of
filters filters preemphasis filter 232 b is balanced by the phase error introduced into the conditioned sum signal by 75μs preemphasis filter 222. Further, in this embodiment, the phase response of staticphase equalization filter 228 is selected to closely match the difference between the phase responses of fixedpreemphasis filter 232 a andfilter 132 b, so that any phase error introduced into the encoded difference signal byfilter 232 a is balanced by a compensatory phase error in the conditioned sum signal that is introduced by staticphase equalization filter 228. -
Clipper 254 performs processing that is partially analogous to the overmodulation protection portion of overmodulation protector and bandlimiter 138 (shown inFIG. 1 ) used inidealized encoder 100. Briefly,clipper 254 is implemented as a thresholding device, however, the operation ofclipper 254 will be discussed in greater detail below. -
Wideband compression unit 280 andspectral compression unit 290 perform processing functions that are partially analogous to that ofunits FIG. 1 ). Briefly,wideband compression unit 280 dynamically compresses the signal online 239 as a function of the overall energy level in the encoded difference signal andspectral compression unit 290 further compresses high frequency portions of the signal online 281 as a function of high frequency energy in the encoded difference signal. -
FIG. 5 shows a block diagram of a preferred embodiment of a digitalwideband compression unit 280.Unit 280 includes adigital signal multiplier 434, adigital signal multiplier 446, a widebanddigital bandpass filter 448, a digitalRMS level detector 450, and a digitalreciprocal generator 458. These components perform processing functions partially analogous to those performed byamplifier 134,amplifier 146,bandpass filter 148,RMS level detector 150, andreciprocal generator 152, respectively, of idealized encoder 100 (shown inFIG. 1 ). The encoded difference signal is applied viafeedback path 240 to an input of widebanddigital bandpass filter 448 which generates therefrom an output signal that is applied toRMS level detector 450. The latter generates an output signal that is representative of the RMS value of the output signal generated byfilter 448 and applies this output signal vialine 450 a toreciprocal generator 458.Reciprocal generator 458 then generates an output signal representative of the reciprocal of the signal online 450 a and applies this output signal vialine 458 a tomultiplier 446.Digital signal multiplier 446 multiplies the signal online 458 a by the value of the gain setting, Gain D, and thereby generates an output signal that is representative of D times the reciprocal of the RMS value and that is applied vialine 446 a to an input terminal ofmultiplier 434. The output signal generated by fixedpreemphasis filter 232 a is applied vialine 239 to another input terminal ofmultiplier 434.Multiplier 434 multiplies the signal online 239 by the signal online 446 a and thereby generates the output ofwideband compression unit 280 which is applied vialine 281 to the input ofspectral compression unit 290. - Wideband
digital bandpass filter 448 is designed to have an amplitude response that closely matches the amplitude response of bandpass filter 148 (shown inFIG. 1 ). One preferred choice is to selectfilter 448 so that the mean square difference between its amplitude response and that offilter 148 are minimised. In one embodiment, the phase response offilters RMS level detector 450 is substantially insensitive to the phase of its input signal, these phase differences maybe ignored. In one preferred embodiment,wideband bandpass filter 448 is implemented as a second order IIR filter having a transfer function H(z) that is described by the formula shown in Equation (4). -
RMS level detector 450 is designed to approximate the performance ofdetector 150 which is used in idealized encoder 100 (shown inFIG. 1 ).Detector 450 includes asignal squaring device 452, asignal averaging device 454, and asquare root device 456. Squaringdevice 452 squares the signal generated bybandpass filter 448 and applies this squared signal vialine 452 a to averagingdevice 454. The latter computes a time weighted average of the signal online 452 a and applies the average vialine 454 a tosquare root device 456.Square root device 456 calculates the square root of the signal online 454 a and thereby generates a signal online 450 a representative of the RMS value of the output signal generated by widebanddigital bandpass filter 448. - Averaging
device 454 includes adigital signal multiplier 460, adigital signal adder 462, adigital signal multiplier 464, and adelay register 465. The output signal generated by squaringdevice 452 is applied vialine 452 a to one input ofmultiplier 460 which generates an output signal by scaling the signal online 452 a by a constant α. The scaled output signal generated bymultiplier 460 is applied to one input ofadder 462 and an output signal generated bydelay register 465 is applied to the other input ofadder 462.Adder 462 generates an output signal by summing the signals present at its two inputs, and this summed signal is the output signal of averagingdevice 454 and is applied tosquare root device 456 vialine 454 a. This summed signal is also applied to one input ofmultiplier 464 which generates an output signal by scaling the summed signal by the constant (1−α). The output signal generated bymultiplier 464 is applied to an input ofdelay register 465. Those skilled in the art will appreciate thataverager 454 is a recursive filter and implements a digital averaging function that is described by the recursive formula shown in the following Equation (5).
y(n)=αx(n)+(1−α)y(n−1) (5)
in which y(n) represents the current digital sample of the signal output byaverager 454 online 454 a, y(n−1) represents the previous digital sample of the signal output byaverager 454 online 454 a, and x(n) represents the current digital sample of the signal output by squaringdevice 452 online 452 a. Those skilled in the art will appreciate thataverager 454 provides a digital approximation of the analog averaging function defined in the BTSC standard and implemented by RMS level detector 150 (shown inFIG. 1 ) ofidealized encoder 100. The constant a is preferably chosen so that the time constant ofRMS level detector 450 closely approximates the corresponding time constant specified in the BTSC standard forRMS level detector 150. - Digital
square root device 456 and digitalreciprocal generator 458 are shown inFIG. 5 as two separate components, however, those skilled in the art will appreciate that these two components may be implemented using a single device that generates an output signal representative of the reciprocal of the square root of its input signal. Such a device may be implemented for example as a memory look up table (LUT), or alternatively may be implemented using processing components that calculate a Taylor series polynomial approximation of the inverse square root function. -
FIG. 6 shows a block diagram of a preferred embodiment ofspectral compression unit 290.Unit 290 includes a variable preemphasis/deemphasis unit (hereinafter referred to as the “variable emphasis unit”) 536, asignal multiplier 540, a spectralband pass filter 542, and anRMS level detector 544, and these components provide processing which is partially analogous to that ofvariable emphasis filter 136,amplifier 140,bandpass filter 142, andRMS level detector 144, respectively, of idealized encoder 100 (shown inFIG. 1 ). The encoded difference signal is applied viafeedback line 240 to an input ofsignal multiplier 540 which generates an output signal by multiplying the encoded difference signal by the fixed gain setting value of Gain C. The amplified output signal generated bysignal multiplier 540 is applied tospectral bandpass filter 542 which generates an output signal that is applied toRMS level detector 544. The latter generates an output signal that is applied vialine 544 a to a control terminal ofvariable emphasis unit 536, and the output signal generated bywideband compressor unit 280 is applied vialine 281 to an input terminal ofunit 536. The latter dynamically varies the frequency response applied to the signal online 281 according to a function of the signal online 544 a, the latter signal being a function of the signal energy of the encoded difference signal within the frequency band passed by spectralband pass filter 542. The output signal ofunit 290, which is generated byunit 536 and is applied to the input of fixedpreemphasis filter 232 b, is thus dynamically compressed a greater amount in the high frequency portions of the signal than in the remainder of the spectrum of interest. -
Spectral bandpass filter 542 is designed to have an amplitude response that closely matches the amplitude response of bandpass filter 142 (shown inFIG. 1 ) ofidealized encoder 100. As with filter 448 (shown inFIG. 5 ), one preferred choice is to selectfilter 542 so that the difference between its RMS amplitude response and that offilter 142 are minimized. In one embodiment, the phase response offilters RMS level detector 544 is substantially insensitive to the phase of the input to the detector, these phase differences may be ignored. In one preferred embodiment,spectral bandpass filter 542 is implemented as a cascade of three second orderIIR filter sections FIG. 6 ) each having a transfer function H(z) that is described by the formula shown in Equation (4). -
RMS level detector 544 is designed to approximate the performance ofdetector 144 which is used in idealized encoder 100 (shown inFIG. 1 ).Detector 544 includes asignal squaring device 552, asignal averaging device 554, and asquare root device 556. Squaringdevice 552 squares the signal generated byspectral bandpass filter 542 and applies this squared signal vialine 552 a to averagingdevice 554. The latter functions similarly to averaging device 454 (shown inFIG. 5 ) which is used in thewideband compression unit 280, althoughdevice 554 preferably uses a constant β different from the constant α. The behavior of averagingdevice 554 is of course also described by Equation (5) when β is substituted for α. The constant β is preferably selected fordevice 554 so that the time constant ofRMS level detector 544 closely approximates the corresponding time constant specified by the BTSC standard for RMS level detector 144 (shown inFIG. 1 ). Averagingdevice 554 computes a time weighted average of the signal online 552 a and applies the average tosquare root device 556 vialine 554 a.Square root device 556 calculates the square root of the signal online 554 a and thereby generates a signal online 544 a as a function of the RMS value of the output signal generated byspectral bandpass filter 542. - The signal on
line 544 a is applied to the control terminal ofvariable emphasis unit 536.Variable emphasis unit 536 performs processing that is partially analogous to filter 136 (shown inFIG. 1 ) ofidealized encoder 100. As defined by the BTSC standard,filter 136 has amplitude and phase responses that vary as a function of the output signal generated byRMS level detector 144. One preferred way to implementunit 536 so that it has similar variable responses is to use a digital filter having variable coefficients that determine its transfer function and to select the value of the coefficients during any given sample period, or group of sample periods, based on the value of the signal online 544 a. -
FIG. 6 shows one embodiment ofvariable emphasis unit 536 which includes alogarithmic generator 558, avariable emphasis filter 560, and a look uptable LUT 562. The output signal generated byRMS level detector 544 is applied vialine 544 a tologarithmic generator 558. The latter generates a signal online 558 a that is representative of the logarithm of the signal online 544 a and applies this signal toLUT 562.LUT 562 generates an output signal selected from the LUT and: representative of filter coefficients to be used byvariable emphasis filter 560. The coefficients thus generated byLUT 562 are applied vialine 562 a to a coefficient selection terminal ofvariable emphasis filter 560. The output signal generated bywideband compression unit 280 is applied to an input terminal ofvariable emphasis filter 560 vialine 281.Variable emphasis filter 560 generates the output signal ofspectral compression unit 290 which is applied to the input of fixedpreemphasis filter 232 b. -
Variable emphasis filter 560 is designed to have a variable amplitude response that closely matches the variable amplitude response of filter 136 (shown inFIG. 1 ) ofidealized encoder 100.Variable emphasis filter 560 provides a similar variable response by using a variable coefficient transfer function (i.e., the coefficients of the transfer function H(z) offilter 560 are variable) and by allowingLUT 562 to select the value of the coefficients during intervals based on the sample period. As will be described in greater detail below,LUT 562 stores the values of the filter coefficients used byfilter 560, and during each sample period, or during any selected group of sample periods,LUT 562 selects a set of filter coefficients as a function of the output signal generated bylogarithmic generator 558 online 558 a. In one preferred embodiment,variable emphasis filter 560 is implemented as a first order IHt filter having a transfer function H(z) that is described by the formula shown in the following Equation (6).
in which the filter coefficients b0, b1, and a1 are variables that are selected byLUT 562. Methods of selecting the values for the filter coefficients used byfilter 560 as well as by the other filters ofencoder 200 will be discussed below. - In
FIG. 6 ,logarithmic generator 558 andsquare root device 556 are shown, for convenience, as two separate components. However, those skilled in the art will appreciate that these two components may be implemented using a single device, such as a LUT, or alternatively using processing components that calculate a Tayler series polynomial approximation of the logarithm of the signal online 554 a and by then dividing this value by two. Similarly, in alternative implementations, the functions performed bylogarithmic generator 558,square root device 556, andLUT 562 maybe incorporated into a single device. - As stated above, high pass filters 212, 214 (shown in
FIG. 3 ) are useful in blocking DC components so as to prevent encoder 200 from exhibiting a behavior known as “ticking”. In the context of a stereophonic encoder, ticking refers to relatively low frequency oscillatory behavior of the encoder caused when there is no signal present at the left and right channel audio inputs. The desired behavior of a stereophonic system when there is no signal present at the audio inputs is to remain silent; however, an encoder connected through a decoder to loudspeakers and exhibiting ticking causes the loudspeakers to emit an audible sound, referred to as a “tick”, with a somewhat regular period that is partially dependent on the time constant of the RMS level detector in the wideband compressor. More particularly, inencoder 200, when only very low level signals are present at the audio inputs, and when there is a D.C. component, or an offset, present in the signal online 239,wideband compression unit 280 tends to behave in an unstable fashion that causes ticking. - Consider the case where only a low level audio signal is present on
line 239. In such a case, the output ofRMS level detector 450 online 450 a becomes very small, which in turn causes the gain ofmultiplier 434 to become very large. If such a low level audio signal online 239 is constant in its amplitude, thewideband compression unit 280 reaches a steady-state condition after some time (determined by the time constant a applied to multiplier 460), because the encoded difference signal is fed back online 240 to thewideband compression unit 280. Because the feedback is arranged to be negative, when the audio signal online 239 increases in its amplitude, the signal online 450 a increases, which in turn causes the gain ofmultiplier 434 to decrease. When the audio signal online 239 decreases in its amplitude, the signal online 450 a decreases, which in turn causes the gain ofmultiplier 434 to increase. - However, should there be a significant dc signal present on
line 239 in addition to a low level audio signal, the dc signal is blocked from the feedback process by the action ofwideband bandpass filter 448, which has zero response to dc signals. In particular, any dc present in the encoded difference signal atline 240 is blocked byfilter 448, and is not sensed byRMS level detector 450. Any dc signal present online 239 will be amplified bymultiplier 434 along with any audio signal present online 239, but the amplification factor or gain will be determined only by the audio signal amplitude as sensed byRMS level detector 450 after filtering byfilter 448. - As noted above, whenever the amplitude of the audio signal on
line 239 varies, the gain ofmultiplier 434 varies inversely. During such variations in gain, any dc present online 239 will also be subjected to variable amplification, in effect modulating the dc signal, thereby producing an ac signal. In this fashion such dc signals may be modulated so as to create significant audio-band signals which will not be rejected byfilter 448, and are therefore sensed bydetector 450. When the audio signal online 239 is small compared to the do online 239, small variations in the audio signal level, which cause changes in the gain ofamplifier 434, can cause a large change in the dc level (which amount to an ac signal) atline 281 through this modulation process. The ac signal produced tends to increase the overall signal which passes throughfilter 448, regardless of whether the audio signal variation that gave rise to the ac signal was an increase or decrease in signal level. In particular, should the level of the audio signal online 239 decrease, the negative feedback process normally increases the gain ofmultiplier 434. However, if a sufficient dc signal is present inline 239, a decrease in audio signal online 239 can cause an increase in the signal sensed bydetector 450, forcing the gain ofmultiplier 434 to decrease. In this fashion, the negative feedback process is reversed, and the feedback becomes positive. - Such positive feedback will only persist so long as the modulated dc signal at
line 281 is sufficiently large compared to any audio signal present online 281, when weighted by the response of all the filters and signal modifiers betweenline 281 and the output offilter 448. Once the gain ofmultiplier 434 decreases sufficiently such that the modulated dc signal inline 281 no longer provides a significant input todetector 450, the feedback reverts to its normal negative sense. In accordance with the time constant ofdetector 450, the system will re-acquire an appropriate gain level based on the level of the audio signal inline 239. But, if sufficient dc remains in the signal inline 239, the cycle will repeat itself once the gain ofmultiplier 434 increases sufficiently. During each such period of positive feedback, a sharp change in the dc level ofline 281 is produced. This change is audible, and sounds somewhat similar to the ‘tick’ of a clock. Since such dc changes will occur with some regularity, based on the time constant ofdetector 450, the phenomenon is often referred to as ‘ticking’. - One method of preventing ticking is to remove any do components present in the input signal to
encoder 200. This is accomplished by high pass filters 212 and 214. Further, high pass filters 212 and 214 help to maximize the dynamic range ofencoder 200 by removing dc components which otherwise may use up valuable dynamic range. - As stated above and as shown in
FIG. 3 , low pass filter 238 is preferably implemented as twofilters filter 238 a were eliminated, and the entire filter 238 were located after clipper 254 (i.e., in the location offilter 238 b) then any components above 15 kHz on the audio input signals may cause instability in thewideband compression unit 280 similar to the above-described ticking behavior. This occurs because any signal components above 15 kHz online 239 will be amplified by multiplier 434 (shown inFIG. 5 ) and because such components will not be sensed byRMS level detector 450 since such components are filtered out by the low pass filter following clipper 254 (shown inFIG. 3 ). Sincedetector 450 increases the gain ofmultiplier 434 when it senses the absence of a signal, the gain ofmultiplier 434 can become relatively large when the signal online 239 consists of little audio signal (under 15 kHz) information, but significant high frequency (over 15 kHz) information.Multiplier 434 then amplifies the high frequency information, which can generate large signals that are likely to be clipped by components inprocessing section 230. This clipping can produce harmonics which may alias to low frequencies that will be sensed byRMS level detector 450 causing the system to tick as described previously. Alternatively, iffilter 238 b were eliminated and the entire filter 238 were located before fixedpreemphasis filter 232 a (i.e., in the location offilter 238 a) then high frequency artifacts generated byclipper 254 would be included in the encoded difference signal and could interfere with the pilot tone in the composite signal. Therefore, splitting filter 238 as shown provides an optimal arrangement wherebyfilter 238 a prevents ticking incompression unit 280 and filter 238 b filters high frequency artifacts that may be generated byclipper 254. - Fixed preemphasis filter 232 is also preferably split into two
filters FIG. 3 . Filter 232 typically requires relatively large gain at high frequencies, as is specified in the BTSC standard, and using only a single section to implement filter 232 increases the likelihood of filter 232 causing clipping. It is advantageous to apply some of the gain of filter 232 on the input side of wideband compression unit 280 (withfilter 232 a) and to apply some of the gain of filter 232 on the output side of wideband compression unit 280 (withfilter 232 b). Sinceunit 280 normally compresses its input signal, distributing the gain of filter 232 around the compression provided byunit 280 decreases that the likelihood that the gain of filter 232 will cause an overflow condition. - To minimize size, power consumption, and cost,
encoder 200 is preferably implemented using a single digital signal processing chip.Encoder 200 has been successfully implemented using one of the well known Motorola DSP 56002 digital signal processing chips (this implementation shall be referred to hereinafter as the “DSP Embodiment”). The Motorola DSP 56002 is a fixed point twenty-four bit chip, however, other types of processing chips, such as floating point chips, or fixed point chips having other word lengths, could of course be used. The DSP Embodiment ofencoder 200, uses a sampling frequency fs that is equal to three tunes the pilot frequency fH (i.e., fs=47202 Hz). The following Table I lists all of the filter coefficients used in the DSP Embodiment ofencoder 200 except those used invariable emphasis filter 560.TABLE 1 Low Pass Filter (Section #1) 310 Low Pass Filter (Section #2) 312 (Equation 4) (Equation 4) b0 = 0.18783270 b0 = 0.44892888 b1 = 0.36310206 b1 = 0.70268024 b2 = 0.18783270 b2 = 0.44892888 a1 = −0.388832539 a1 = 0.12638618 a2 = 0.12709286 a2 = 0.47415181 Low Pass Filter (Section #3) 314 Low Pass Filter (Section #4) 316 (Equation 4) (Equation 4) b0 = 0.70674027 b0 = 0.85733126 b1 = 0.87637648 b1 = 0.91505047 b2 = 0.70674027 b2 = 0.85733126 a1 = 0.53702472 a1 = 0.74320197 a2 = 0.75298490 a2 = 0.89832289 Low Pass Filter (Section #5) 318 Wideband Bandpass Filter 448 (Equation 4) (Equation 4) b0 = 0.92737972 b0 = −0.02854672 b1 = 0.92729649 b1 = −0.18789051 b2 = 0.92737972 b2 = 0.21643723 a1 = 0.82951974 a1 = −1.75073141 a2 = 0.97259237 a2 = 0.75188028 Fixed Preemphasis Filter 238a Fixed Preemphasis Filter 238b (Equation 2) (Equation 2) b0 = 9.50682180 b0 = 4.357528 b1 = 9.00385663 b1 = −3.24843271 a1 = −0.497064357 a1 = 0.10881833 Spectral Bandpass Filter Spectral Bandpass Filter (Section #3) 542a (Section #2) 542b (Equation 4) (Equation 4) b0 = 0.646517841 b0 = 0.850281278 b1 = 0.649137616 b1 = −0.850247036 b2 = 0.0 b2 = 0.0 a1 = 0.557821757 a1 = −0.602159890 a2 = 0.0 a2 = 0.0 Spectral Bandpass Filter Static Phase Equalization Filter (Section #3) 542c 224 (Equation 4) (Equation 3) b0 = 0.597678418 a0 = 0.9029 b1 = −1.195357770 b2 = 0.597679348 a1 = −0.776566094 a2 = 0.352824276 75 μs preemphasis filter 222 High Pass Filters 212, 214 (Equation 2) (Equation 1) b0 = 4.57030583 a1 = −0.999 b1 = −3.43823487 a1 = 0.131778883 - In the DSP Embodiment of
encoder 200 the value of the constant a that is used by averager 454 (shown inFIG. 5 ) inwideband compression unit 280 is set equal to 0.0006093973517, and the value of the constant β that is used by averager 554 (shown inFIG. 6 ) inspectral compression unit 290 is set equal to 0.001825967. Further, the values of Gain C and Gain D used byamplifiers encoder 200 performs similarly toencoder 100. -
FIG. 7 shows aflow chart 700 that describes one preferred method for pre-calculating all the sets of filter coefficients used by variable emphasis filter 560 (shown inFIG. 6 ) in the DSP Embodiment ofencoder 200. Prior to operation ofencoder 200, all the sets of filter coefficients used byfilter 560 are pre-calculated (e.g., by a general purpose digital computer) and are loaded intoLUT 562. In the DSP Embodiment ofencoder 200,filter 560 has a transfer function H(z) that is described by Equation (6) soflow chart 700 describes the calculation of the coefficients b0, b1, and a1. As specified in the BTSC standard, the transfer function of S(f,b) of analog filter 136 (shown inFIG. 1 ) to whichfilter 560 partially corresponds, is described by the formula shown in the following Equation (7). - in which F is equal to 20.1 kHz.
- The first step in
flow chart 700 in aninitialization step 710 during which several variables are initialized. Specifically, the sampling frequency fs is set equal to 47202 Hz, and the period T is set equal to 1/fs. The variable W is a digital version of the variable F used in Equation (7) and is set equal to π(20.1 kHz)/fs. The variable dBRANGE represents the desired signal range of the RMS detectors in the spectral compression unit, and for the DSP Embodiment ORANGE is set equal to 72.25 M. The variable dBRF.S relates to the sensitivity offilter 560 to changes in the energy level of the encoded difference signal. In the DSP Embodiment ofencoder 200, dBRES is set equal to 0.094 dB so thatfilter 560 will use coefficients based on the value of the signal online 558 a quantized to the nearest 0.094 db. The variable N equals the total number of sets of filter coefficients used infilter 560 and N is calculated by dividing the sensitivity (dBRES) into the range (dBRANGE) and rounding to the nearest integer. In the DSP Embodiment, N is equal to 768 although those skilled in the art will appreciate that this number can be changed which will vary the sensitivity or the range. In the DSP Embodiment,LUT 562 stores 769 sets of coefficients forfilter 560, and of course if N is increased, a larger LUT will be used to store the extra sets of filter coefficients. Further, those skilled in the art will appreciate thatlogarithmic generator 558 scales the signal online 558 a and thereby reduces the number of filter coefficient sets stored byLUT 562, for a given minimum quantization of the value of the signal onLine 558 a. However, in other embodiments,logarithmic generator 558 may be eliminated andLUT 562 may store a correspondingly larger number of filter coefficient sets. Finally, the variables Scale and Address are set equal to 32 and zero, respectively. The variable Scale, which is only used in fixed point implementations, is selected so that all the filter coefficients have a value greater than or equal to negative one and less than one (where the filter coefficients are represented in twos complement). - Following
initialization step 710, acoefficient generation step 720 is executed. During the first execution ofstep 720, variables b0(0), b1(0), and a1(0) are calculated which correspond to values of the coefficients bo, b1, and a1 that are to be stored at address location zero ofLUT 562. Following this execution ofstep 720, an incrementingstep 730 is executed during which the value of the variable Address is incremented. Following step 730 a comparison step is executed during which the values of the variables Address and N are compared. If Address is less than or equal to N, then steps 720, 730, and 740 are reexecuted iteratively so that values of the coefficients bo, b1, and a1 are calculated for each of the 769 addresses ofLUT 562. Whenstep 740 detects that the value of Address is greater than N, then all 769 sets of coefficients have been calculated and execution offlow chart 700 proceeds to a concludingstep 750. - In
coefficient generation step 720, the variable dBFS corresponds to the output oflogarithmic generator 558. As the value of the variable Address ranges from zero to 769, the value of dBFS ranges from about −72.25 to zero dB corresponding to the signal range of about 72.25 dB provided by the DSP Embodiment of encoder 200 (where zero dB corresponds to the full modulation). The variable RMSd corresponds to the output of the analog RMS level detector 144 (shown inFIG. 1 ), and as the variable Address ranges from zero to 769, the value of RMSd ranges from about −36 to 36 dB corresponding to the signal range of 72 dB provided by typical prior art analog BTSC encoders. The variable RMSb is a linear version of the variable RMSd, and RMSb corresponds to the variable b in the transfer function S(f,b) described in Equation M. The variables KI and K2 correspond to the (b+51)/(b+1) and the (51b+1)/(0+1) terms, respectively, in Equation (7). The coefficients b0, b1, and a1 are calculated as shown instep 720 using the variables K1; K2, W, and Scale. -
FIG. 8A shows a block diagram that illustrates one method of using the DSP Embodiment in an analog system, and inFIG. 8A , all components that are implemented in the 56002 integrated circuit are indicated at 200 a. The analog system supplies analog left and right channel audio input signals (shown inFIG. 8A as “L” and “R”, respectively) and these signals are applied to the inputs of sixteen bit analog-to-digital converters Converters converters converters modules Modules modules - As stated above, the 56002 chip is a fixed point twenty-four bit processor, and the samples applied to the chip by
converters Modules converters modules converters encoder 200 a to exceed sixteen bits without causing an error condition such as an overflow. - In
encoder 200 a, each bit of the twenty-four bit word corresponds roughly to 6 dB of signal range, and thereforemodules converters modules -
Input section 210 receives the twenty-four bit words generated bymodules channel processing section 220. The output signal generated by sumchannel processing section 220 is applied to a “times 16 module” (which may be considered as a 24 dB amplifier) 296.Module 296 thereby compensates for the −24dB attenuators channel processing section 220 back to 100% modulation (i.e., back to “full scale”). The output signal generated bymodule 296 is applied to a sixteen bit digital-toanalog converter 814 which in turn generates an analog conditioned sum signal. -
Input section 210 also generates the difference signal that is applied to the differencechannel processing section 230. As stated above, as a result ofmodules encoder 200 a, clipper 254 (shown inFIG. 3 ) of thedifference processing section 230 includes an 18 dB amplifier (which is implemented as a multiply by eight). That is,clipper 254 amplifies the signal generated by fixedpreemphasis filter 232 b by 18 dB and then clips this amplified signal so that the output signal generated byclipper 254 will not exceed a number that is 6 dB down from full modulation. The signal applied fromclipper 254 tolow pass filter 238 b therefore has one bit (or 6 dB) of “headroom”, so filter 238 b may generate an output signal that is 6 dB greater than its input signal without causing saturation. It is desirable to leave this one bit of headroom because the transient response offilter 238 b includes some ringing that may cause it to temporarily generate an instantaneous output signal that is greater than its instantaneous input signal and the headroom thereby prevents any ringing infilter 238 b from causing a saturation condition. Referring again toFIG. 8A , the output signal generated byfilter 238 b is applied to a sixteen bit digital-to-analog converter 816 which in turn generates an output signal that is applied to a 6dB analog amplifier 820. Both D/A converters Converters Amplifier 820 amplifies its input signal by 6 dB and thereby brings the encoded difference signal back up to full scale. WhileFIG. 8A showsencoder 200 a coupled to analog-to-digital converters digital systems converters encoder 200 a receives the digital audio signals directly. -
FIG. 8B shows a block diagram of one preferred embodiment of aBTSC encoder 200 b constructed according to the invention and configured as part of an analog system.Encoder 200 b is similar toencoder 200 a, however, inencoder 200b module 296 amplifies its input signal by 18 dB (by multiplying by 8) rather than by 24 dB as inencoder 200 a. The output signal generated bymodule 296 is a scaled version of the conditioned sum signal and is shown inFIG. 8B as S. Also,encoder 200 b includes amodule 298 for amplifying the output signal generated by differencechannel processing section 230 by 6 dB (by multiplying by two). The output signal generated bymodule 298 is a scaled version of the encoded difference signal and is shown inFIG. 8B as D. Further,encoder 200 b includes acomposite modulator 822 for receiving the signals S and D and for generating therefrom a digital version of the composite signal. The digital composite signal generated bymodulator 822 is applied to a digital-to-analog converter 818 the output of which is an analog version of the composite signal. D/A converter 818 is intended to be a complete converter which includes the aforementioned analog anti-image filter as part of its functionality. Such converters are commonly available in commercial embodiments, such as the Burr-Brown PCM1710. In the preferred embodiments,modules input section 210, sumchannel processing section 220, differencechannel processing section 230,modules composite modulator 822 are all implemented on a single digital signal processing chip. - Since the composite signal is generated as a digital signal in
encoder 200 b,module 298 is included to bring the output signal generated by differencechannel processing section 230 up to full scale rather than waiting until after digital-to-analog conversion and using an analog amplifier such asamplifier 820 as is shown inFIG. 8A . Also, since in the composite signal the conditioned sum signal is used at 50% modulation,module 296 only amplifies its input signal by 18 dB so that the output signal generated bymodule 296 is at half the amplitude of the output signal generated bymodule 298. -
FIG. 9 shows a block diagram of one embodiment ofcomposite modulator 822. The latter receives the signals S and D and generates therefrom a digital version of the composite signal.Modulator 822 includes twointerpolators digital signal multiplier 918, and twodigital signal adders interpolators Interpolators interpolators interpolators filter 916 is applied to one input ofsignal multiplier 918 and a digital oscillating signal as a function of cos [4π(fH/fs)n] is applied to the other input ofmultiplier 918.Multiplier 918 thereby generates the amplitude modulated, double-sideband, suppressed carrier version of the difference signal that is used in the composite signal. The output signal generated bymultiplier 918 is applied to one input ofsignal adder 920 and the filtered output signal generated byfilter 914 is applied to the other input ofsignal adder 920. The latter generates an output signal by summing the two signals present at its inputs and applies this signal to signaladder 922. A pilot tone signal that oscillates as a function of A cos [2π(fH/fs)n] (where ‘A’ is a constant representative of 10% of full scale modulation) is applied to the other input ofsignal adder 922 which generates the digital composite signal by summing the two signals present at its inputs. -
Composite modulator 822 includesinterpolators FIG. 2 ), and therefore the signals applied to the inputs ofsignal multiplier 918 andsignal adder 920 should have sample rates at least as large as 6fH to satisfy the Nyquist criteria. Because the sample rate at the output ofcomposite modulator 822 is typically higher than the sample rate of either the S or D signals, D/Aconverter 818 must be capable of operating at such higher sample rates. If the input signals S and D applied tocomposite modulator 822 have sample rates of 3fH some form of interpolation (such as that provided byinterpolators 910, 912) should be provided to double the sample rate. Of course, if sufficiently high sample rates are used throughoutencoder 200 b then interpolators 910, 912 and low pass filters 914, 916 may be eliminated frommodulator 822. -
FIG. 8C shows a block diagram of yet another embodiment of aBTSC encoder 200 c constructed according to the invention.Encoder 200 c is similar toencoder 200 b (shown inFIG. 8B ) however, inencoder 200c module 298 is eliminated so that the signal generated by the differencechannel processing section 230 is the signal D and is applied directly tocomposite modulator 822. Further, inencoder 200 c,module 296 amplifies its input signal by 12 dB (by multiplying by 4) rather than by 18 dB as is done inencoder 200 b. So inencoder 200 c, the signals S and D are 6 dB down from the levels of those signals inencoder 200 b.Composite modulator 822 therefore generates from these signals a version of the composite signal that is attenuated by 6 dB. This attenuated version of the composite signal is converted to an analog signal by digital-to-analog converter 818 and is then brought up to full scale by 6dB analog amplifier 820. As withencoder 200 b,encoder 200 c is preferably implemented using a single digital signal processing chip. - The differences between
encoders Encoder 200 b minimizes the loss of signal-to-noise ratio as a result of the operation ofconverter 818 by usingmodules converter 818 is at full scale. However, althoughconverter 200 b minimizes any loss of signal-to-noise ratio that might occur as a result ofconverter 818,encoder 200 b also increases the likelihood that clipping might occur in the composite signal. Since the differencechannel processing section 230 uses the relatively large gain provided by fixed preemphasis filter 232 (shown inFIG. 3 ), it is possible for some clipping to occur in the path of the encoded difference signal.Encoder 200 b usesmodule 298 to bring the D signal up to full scale and this essentially eliminates any headroom from the signal path of the D signal and thereby increases the chance that some clipping will occur. So encoder 200 b minimizes the loss of any signal-to-noise ratio that occurs as a result ofconverter 818 at the cost of increasing the likelihood of clipping in the path of the encoded difference signal. In contrast,encoder 200 c preserves headroom in the path of the encoded difference signal and thereby reduces the likelihood of clipping at the cost of increasing the loss of signal-to-noise ratio that occurs as a result of operation ofconverter 818. -
FIG. 8D shows a block diagram of yet another embodiment of aBTSC encoder 200 d constructed according to the invention.Encoder 200 d is similar toencoder 200 a (shown inFIG. 8A ) however,encoder 200 d additionally includes aportion 822 a of a composite modulator.Portion 822 a includes twointerpolators digital signal multiplier 918 and a digital signal adder 930. The S signal generated bymodule 296 is applied tointerpolator 910 which “up-samples” the S signal and applies the up-sampled signal tolow pass filter 914. The latter filters this signal and applies the filtered signal to one input terminal of adder 930. A digital pilot tone having twice the normal amplitude (i.e., 2A cos 2π(fH/fs)n) is applied to the other input terminal of adder 930 which generates an output signal by summing the two signals present at its input terminals. The D signal generated by differencechannel processing section 230 is applied tointerpolator 912 which generates an up-sampled signal that is applied tolow pass filter 916. The latter filters this signal and applies the filtered signal to one terminal ofmultiplier 918. A signal oscillating according to cos 4π(fH/fs)n is applied to the other terminal ofmultiplier 918 which generates an output signal by multiplying the two signals present at its input terminals. As withencoders 200 a-c,encoder 200 d is preferably implemented using a single digital signal processing chip. -
Encoder 200 d is preferably used in conjunction with two digital-to-analog converters dB attenuator 936, an analog 6dB amplifier 938, and ananalog adder 940. The output signal generated by adder 930 is applied toconverter 932 which generates an analog signal that is applied toattenuator 936. The output signal generated bymultiplier 918 is applied toconverter 934 which generates an analog signal that is applied toamplifier 938. The signals generated byattenuator 936 andamplifier 938 are applied to input terminals ofsignal adder 940 which sums these signals to generate the analog composite signal. D/A converters Converters - It is also possible to eliminate
interpolator 910 andlow pass filter 914 fromFIG. 8D , and run D/A converter 932 at a sample rate equal to that of the sumchannel processing section 220. However, to do so is generally not practical because inexpensive, commonly available D/A converters are usually available in pairs housed within a single integrated circuit. Such paired D/A converters naturally operate at the same sample rate. While it is possible to reduce DSP complexity by eliminatinginterpolator 910 andlow pass filter 914 fromFIG. 8D , doing so would also likely increase the cost and complexity of the overall design because a simple stereo D/A converter could no longer be used for both D/A converters -
Encoder 200 d represents one combination of the features ofencoders Encoder 200 d usesmodule 296 to bring the S signal up to full scale so as to minimise any loss of signal-to-noise ratio that might occur as a result of the operation ofconverter 932.Encoder 200 d also preserves 6 dB of headroom in the signal path of the D signal and therefore reduces the likelihood of any loss of accuracy due to clipping. Althoughencoder 200 d includes more components than either ofencoders encoder 200 d both minimizes loss of signal-to-noise ratio and the likelihood of clipping. -
FIG. 10 shows a block diagram of a preferred embodiment of sumchannel processing section 220 a and differencechannel processing section 230 a for use in encoder 200 (and thesesections encoders 200 a-d). Processingsections sections section 220 a additionally includes dynamicphase equalization filter 1010, andsection 230 a additionally includes a dynamicphase equalization filter 1012. In the illustrated embodiment, the output signals generated by staticphase equalization filter 228 and fixedpreemphasis filter 232 a are applied to the input terminals of dynamicphase equalization filters logarithmic generator 558 online 558 a is applied to the control terminals ofMn filters low pass filter 224 and towideband compression unit 280, respectively. - Dynamic
phase equalization filters variable emphasis filter 560 which is used inspectral compression unit 290. The phase response ofvariable emphasis filter 560 is preferably matched as closely as is possible to that of variable emphasis filter 136 (shown inFIG. 1 ). However, due to the variable, signal dependent, nature ofvariable emphasis filter 136, it is extremely difficult to designvariable emphasis filter 560 so that its phase response is matched to that ofvariable emphasis filter 136 for all pre-emphasis/de-emphasis characteristics, which in turn varies with signal level. Therefore in typical embodiments ofencoder 200, the phase responses ofvariable emphasis filter 560 andvariable emphasis filter 136 diverge as a function of the signal level. Dynamicphase equalization filters variable emphasis filter 560 andvariable emphasis filter 136. - Dynamic
phase equalization filters phase equalization filter 228. However, whereasfilter 228 compensates for phase errors that are independent of the level of the encoded difference signal, filters 1010, 1012 compensate for phase errors that are dependent on this signal level.Filters variable emphasis filter 560. In preferred embodiments,filters variable emphasis unit 536 and include a filter having a variable coefficient transfer function and a LUT for selecting the values of the filter coefficients during any particular interval. The signal generated bylogarithmic generator 558 online 558 a is preferably applied to the control terminals offilters -
Digital encoder 200 has been discussed in connection with certain particular embodiments, however, those skilled in the art will appreciate that variations of these embodiments are also embraced within the invention. For example, variable emphasis unit 536 (shown inFIG. 6 ) has been discussed in terms of being implemented using avariable emphasis filter 560 and aLUT 562. However, rather than precomputing all the possible coefficients forfilter 560 and storing them inLUT 562, it may be preferable for other implementations ofvariable emphasis unit 536 to eliminateLUT 562 and to instead include components for calculating the filter coefficients in real time. Those skilled in the art will appreciate that such considerations represent a tradeoff between memory resources (such as are used by a LUT for storing filter coefficients) and computing resources (such as are used by components for calculating filter coefficients in real time) and may be resolved differently in any particular implementation ofencoder 200. Similar considerations apply tosquare root devices reciprocal generator 458, and logarithmic generator 558 (shown inFIGS. 5 and 6 ) which may alternatively use memory resources (e.g., a LUT for storing all the values) or processing resources (e.g., for calculating a Taylor series polynomial approximation). In yet other embodiments, any or all of the components inencoder 200 may be implemented using individual hardware components or alternatively as software modules running on a general or specific purpose computer. - Another example of variations of
encoder 200 that are embraced within the invention relates to scalingmodules 292, 294 (shown inFIG. 8B ). The modules are particularly relevant to fixed point implementations ofencoder 200. In floating point implementations there is no need to pad each sample with zeros and sign bits to prevent overflow and these modules can therefore be eliminated from floating point implementations. As a further example, the static phase equalization filter 228 (shown inFIG. 10 ) has been discussed in terms of compensating for phase errors introduced byfilter 232 a, however, filter 228 may be alternatively used to compensate for other phase errors introduced by other components in the differencechannel processing section 230 a. Still further,filters - Therefore, since certain changes may be made in the above apparatus without departing from the scope of the invention herein involved, it is intended that all matter contained in the above description or shown in the accompanying drawing shall be interpreted in an illustrative and not a limiting sense.
Claims (1)
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Families Citing this family (100)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US5796842A (en) * | 1996-06-07 | 1998-08-18 | That Corporation | BTSC encoder |
US8908872B2 (en) * | 1996-06-07 | 2014-12-09 | That Corporation | BTSC encoder |
US5953067A (en) * | 1997-02-10 | 1999-09-14 | Cable Electronics, Inc. | Multichannel television sound stereo and surround sound encoder |
US6288747B1 (en) * | 1997-08-25 | 2001-09-11 | Cable Electronics, Inc. | Multichannel television sound stereo and surround sound encoder suitable for use with video signals encoded in plural formats |
US6463410B1 (en) * | 1998-10-13 | 2002-10-08 | Victor Company Of Japan, Ltd. | Audio signal processing apparatus |
US6757396B1 (en) | 1998-11-16 | 2004-06-29 | Texas Instruments Incorporated | Digital audio dynamic range compressor and method |
US6192086B1 (en) * | 1999-01-14 | 2001-02-20 | Antec Corporation | Digital sub-systems and building blocks for a mostly digital low-cost BTSC compatible encoder |
US7397850B2 (en) * | 1999-02-18 | 2008-07-08 | Easley Mathew F | Reciprocal index lookup for BTSC compatible coefficients |
US6281813B1 (en) * | 1999-07-09 | 2001-08-28 | Micronas Gmbh | Circuit for decoding an analog audio signal |
JP3693230B2 (en) * | 1999-12-27 | 2005-09-07 | 株式会社エヌ・ティ・ティ・ドコモ | Packet communication system |
WO2004100604A1 (en) * | 2000-02-18 | 2004-11-18 | Arris International, Inc. | Reciprocal index lookup for btsc compatible coefficients |
US6618486B2 (en) * | 2000-05-03 | 2003-09-09 | Robert A. Orban | Controller for FM 412 multiplex power regulation |
US8019091B2 (en) | 2000-07-19 | 2011-09-13 | Aliphcom, Inc. | Voice activity detector (VAD) -based multiple-microphone acoustic noise suppression |
US8280072B2 (en) | 2003-03-27 | 2012-10-02 | Aliphcom, Inc. | Microphone array with rear venting |
US6577189B2 (en) * | 2000-07-20 | 2003-06-10 | Tripath Technology, Inc. | Scheme for reducing transmit-band noise floor and adjacent channel power with power backoff |
DE10116358A1 (en) * | 2001-04-02 | 2002-11-07 | Micronas Gmbh | Device and method for the detection and suppression of faults |
US7046750B1 (en) * | 2001-04-09 | 2006-05-16 | Micronas Gmbh | Adaptive signal weighting system |
DE10124699C1 (en) * | 2001-05-18 | 2002-12-19 | Micronas Gmbh | Circuit arrangement for improving the intelligibility of speech-containing audio signals |
US8452023B2 (en) * | 2007-05-25 | 2013-05-28 | Aliphcom | Wind suppression/replacement component for use with electronic systems |
US6954534B2 (en) * | 2001-07-11 | 2005-10-11 | Kima Wireless Technologies, Inc. | Multiple signal carrier transmission apparatus and method |
US6832078B2 (en) * | 2002-02-26 | 2004-12-14 | Broadcom Corporation | Scaling adjustment using pilot signal |
US6859238B2 (en) * | 2002-02-26 | 2005-02-22 | Broadcom Corporation | Scaling adjustment to enhance stereo separation |
US7079657B2 (en) * | 2002-02-26 | 2006-07-18 | Broadcom Corporation | System and method of performing digital multi-channel audio signal decoding |
DE10255687B4 (en) * | 2002-11-28 | 2011-08-11 | Lantiq Deutschland GmbH, 85579 | Method for reducing the crest factor of a multi-carrier signal |
WO2004051627A1 (en) * | 2002-11-29 | 2004-06-17 | Koninklijke Philips Electronics N.V. | Audio coding |
US9066186B2 (en) | 2003-01-30 | 2015-06-23 | Aliphcom | Light-based detection for acoustic applications |
EP2254351A3 (en) * | 2003-03-03 | 2014-08-13 | Phonak AG | Method for manufacturing acoustical devices and for reducing wind disturbances |
US7215705B2 (en) * | 2003-03-17 | 2007-05-08 | Intel Corporation | Reducing phase noise in phase-encoded communications signals |
US9099094B2 (en) | 2003-03-27 | 2015-08-04 | Aliphcom | Microphone array with rear venting |
CN1795697A (en) * | 2003-04-09 | 2006-06-28 | 塔特公司 | Reciprocal index lookup for BTSC compatible coefficients |
US7132874B2 (en) * | 2003-04-23 | 2006-11-07 | The Regents Of The University Of Michigan | Linearizing apparatus and method |
US7369603B2 (en) * | 2003-05-28 | 2008-05-06 | Intel Corporation | Compensating for spectral attenuation |
RU2347282C2 (en) * | 2003-07-07 | 2009-02-20 | Конинклейке Филипс Электроникс Н.В. | System and method of sound signal processing |
US7532728B2 (en) * | 2003-08-14 | 2009-05-12 | Broadcom Corporation | Mechanism for using the allpass decomposition architecture for the cauer low pass filter used in a BTSC |
US7557862B2 (en) * | 2003-08-14 | 2009-07-07 | Broadcom Corporation | Integrated circuit BTSC encoder |
SE527670C2 (en) * | 2003-12-19 | 2006-05-09 | Ericsson Telefon Ab L M | Natural fidelity optimized coding with variable frame length |
US7809579B2 (en) | 2003-12-19 | 2010-10-05 | Telefonaktiebolaget Lm Ericsson (Publ) | Fidelity-optimized variable frame length encoding |
US7403624B2 (en) * | 2003-12-23 | 2008-07-22 | Freescale Semiconductor, Inc. | BTSC encoder and integrated circuit |
US8054993B1 (en) | 2004-03-13 | 2011-11-08 | Harman International Industries, Incorporated | System for automatic compensation of low frequency audio based on human loudness perceptual models |
WO2005094529A2 (en) * | 2004-03-24 | 2005-10-13 | That Corporation | Configurable filter for processing television audio signals |
WO2005122410A1 (en) * | 2004-06-10 | 2005-12-22 | Koninklijke Philips Electronics N.V. | Method of cyclically converting an analog signal to a multi-bit digital output signal and converter for performing the method |
JP2008509434A (en) | 2004-08-03 | 2008-03-27 | ザット コーポレーション | Upsampling of television audio signals for encoding |
US10848118B2 (en) | 2004-08-10 | 2020-11-24 | Bongiovi Acoustics Llc | System and method for digital signal processing |
US10158337B2 (en) | 2004-08-10 | 2018-12-18 | Bongiovi Acoustics Llc | System and method for digital signal processing |
US8284955B2 (en) | 2006-02-07 | 2012-10-09 | Bongiovi Acoustics Llc | System and method for digital signal processing |
US11431312B2 (en) | 2004-08-10 | 2022-08-30 | Bongiovi Acoustics Llc | System and method for digital signal processing |
KR101335359B1 (en) * | 2004-08-17 | 2013-12-03 | 댓 코포레이션 | Configurable recursive digital filter for processing television audio signals |
KR100616618B1 (en) * | 2004-09-03 | 2006-08-28 | 삼성전기주식회사 | RF modulator |
US7719616B2 (en) * | 2004-09-17 | 2010-05-18 | That Corporation | Direct digital encoding and radio frequency modulation for broadcast television application |
US20060083384A1 (en) * | 2004-10-15 | 2006-04-20 | Luciano Zoso | Amplitude and phase compensator for BTSC encoder |
US20060188102A1 (en) * | 2005-02-18 | 2006-08-24 | Luciano Zoso | BTSC encoding method with digital FM modulation |
US20060188103A1 (en) * | 2005-02-18 | 2006-08-24 | Luciano Zoso | BTSC encoder with digital FM modulator feature |
US9626973B2 (en) | 2005-02-23 | 2017-04-18 | Telefonaktiebolaget L M Ericsson (Publ) | Adaptive bit allocation for multi-channel audio encoding |
US7619639B1 (en) * | 2005-09-12 | 2009-11-17 | Nvidia Corporation | Adaptive scaling using a programmable video engine |
US7774079B2 (en) * | 2005-10-03 | 2010-08-10 | Sigmatel, Inc. | Method and system for receiving and decoding audio signals |
US8200699B2 (en) * | 2005-12-01 | 2012-06-12 | Microsoft Corporation | Secured and filtered personal information publishing |
US10848867B2 (en) | 2006-02-07 | 2020-11-24 | Bongiovi Acoustics Llc | System and method for digital signal processing |
US10701505B2 (en) | 2006-02-07 | 2020-06-30 | Bongiovi Acoustics Llc. | System, method, and apparatus for generating and digitally processing a head related audio transfer function |
US10069471B2 (en) | 2006-02-07 | 2018-09-04 | Bongiovi Acoustics Llc | System and method for digital signal processing |
US7945058B2 (en) * | 2006-07-27 | 2011-05-17 | Himax Technologies Limited | Noise reduction system |
JP4940888B2 (en) * | 2006-10-23 | 2012-05-30 | ソニー株式会社 | Audio signal expansion and compression apparatus and method |
JP2008206136A (en) * | 2007-01-23 | 2008-09-04 | Rohm Co Ltd | Filter circuit, fm transmitter including the same, and electronic equipment using filter circuit and fm transmitter |
US8391507B2 (en) * | 2008-08-22 | 2013-03-05 | Qualcomm Incorporated | Systems, methods, and apparatus for detection of uncorrelated component |
JP5419413B2 (en) * | 2008-10-10 | 2014-02-19 | シャープ株式会社 | Transmission / reception device, transmission / reception method, and program |
US8433578B2 (en) * | 2009-11-30 | 2013-04-30 | At&T Intellectual Property I, L.P. | System and method for automatically generating a dialog manager |
US8897351B2 (en) * | 2010-09-23 | 2014-11-25 | Intel Corporation | Method for peak to average power ratio reduction |
US8670572B2 (en) * | 2010-11-19 | 2014-03-11 | Fortemedia, Inc. | Analog-to-digital converter and analog-to-digital conversion method |
US9420394B2 (en) | 2011-02-16 | 2016-08-16 | Apple Inc. | Panning presets |
US8654984B2 (en) * | 2011-04-26 | 2014-02-18 | Skype | Processing stereophonic audio signals |
US9059786B2 (en) * | 2011-07-07 | 2015-06-16 | Vecima Networks Inc. | Ingress suppression for communication systems |
US8965774B2 (en) * | 2011-08-23 | 2015-02-24 | Apple Inc. | Automatic detection of audio compression parameters |
EP2709106A1 (en) * | 2012-09-17 | 2014-03-19 | Fraunhofer-Gesellschaft zur Förderung der angewandten Forschung e.V. | Apparatus and method for generating a bandwidth extended signal from a bandwidth limited audio signal |
EP2757558A1 (en) * | 2013-01-18 | 2014-07-23 | Fraunhofer-Gesellschaft zur Förderung der angewandten Forschung e.V. | Time domain level adjustment for audio signal decoding or encoding |
US9398394B2 (en) * | 2013-06-12 | 2016-07-19 | Bongiovi Acoustics Llc | System and method for stereo field enhancement in two-channel audio systems |
US9883318B2 (en) | 2013-06-12 | 2018-01-30 | Bongiovi Acoustics Llc | System and method for stereo field enhancement in two-channel audio systems |
US9264004B2 (en) | 2013-06-12 | 2016-02-16 | Bongiovi Acoustics Llc | System and method for narrow bandwidth digital signal processing |
US9906858B2 (en) | 2013-10-22 | 2018-02-27 | Bongiovi Acoustics Llc | System and method for digital signal processing |
US10820883B2 (en) | 2014-04-16 | 2020-11-03 | Bongiovi Acoustics Llc | Noise reduction assembly for auscultation of a body |
US10639000B2 (en) | 2014-04-16 | 2020-05-05 | Bongiovi Acoustics Llc | Device for wide-band auscultation |
US9615813B2 (en) | 2014-04-16 | 2017-04-11 | Bongiovi Acoustics Llc. | Device for wide-band auscultation |
US9638672B2 (en) | 2015-03-06 | 2017-05-02 | Bongiovi Acoustics Llc | System and method for acquiring acoustic information from a resonating body |
JP2018530451A (en) * | 2015-07-31 | 2018-10-18 | ハイアー ディメンション マテリアルズ,インコーポレイティド | Embossed fabric assembly |
EP3334031B1 (en) * | 2015-08-05 | 2021-05-05 | Panasonic Intellectual Property Management Co., Ltd. | Motor control device |
US9621994B1 (en) | 2015-11-16 | 2017-04-11 | Bongiovi Acoustics Llc | Surface acoustic transducer |
JP2018537910A (en) | 2015-11-16 | 2018-12-20 | ボンジョビ アコースティックス リミテッド ライアビリティー カンパニー | Surface acoustic transducer |
US11129596B2 (en) * | 2016-10-06 | 2021-09-28 | General Electric Company | Systems and methods for ultrasound multiplexing |
JP7076824B2 (en) * | 2017-01-04 | 2022-05-30 | ザット コーポレイション | System that can be configured for multiple audio enhancement modes |
US11245375B2 (en) | 2017-01-04 | 2022-02-08 | That Corporation | System for configuration and status reporting of audio processing in TV sets |
MX2020010166A (en) * | 2018-03-28 | 2020-10-22 | That Corp | System for configuration and status reporting of audio processing in tv sets. |
KR20200143707A (en) | 2018-04-11 | 2020-12-24 | 본지오비 어커스틱스 엘엘씨 | Audio enhancement hearing protection system |
US10959035B2 (en) | 2018-08-02 | 2021-03-23 | Bongiovi Acoustics Llc | System, method, and apparatus for generating and digitally processing a head related audio transfer function |
US10574256B1 (en) * | 2018-09-25 | 2020-02-25 | Cirrus Logic, Inc. | Modulators |
US10819363B2 (en) | 2018-09-25 | 2020-10-27 | Cirrus Logic, Inc. | Modulators |
EP3928315A4 (en) * | 2019-03-14 | 2022-11-30 | Boomcloud 360, Inc. | Spatially aware multiband compression system with priority |
JP7267461B2 (en) | 2019-05-10 | 2023-05-01 | 北京字節跳動網絡技術有限公司 | Video data processing method, apparatus, storage medium and storage method |
EP3967032A4 (en) | 2019-06-07 | 2022-07-27 | Beijing Bytedance Network Technology Co., Ltd. | Conditional signaling of reduced secondary transform in video bitstreams |
CN114208183A (en) | 2019-08-03 | 2022-03-18 | 北京字节跳动网络技术有限公司 | Position-based mode derivation in reduced quadratic transforms of video |
WO2021032045A1 (en) | 2019-08-17 | 2021-02-25 | Beijing Bytedance Network Technology Co., Ltd. | Context modeling of side information for reduced secondary transforms in video |
US10904690B1 (en) | 2019-12-15 | 2021-01-26 | Nuvoton Technology Corporation | Energy and phase correlated audio channels mixer |
KR20210146132A (en) * | 2020-05-26 | 2021-12-03 | 삼성전자주식회사 | Method for calibrating the characteristics of a microphone and electronic device thereof |
Citations (39)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US3757047A (en) * | 1970-05-21 | 1973-09-04 | Sansui Electric Co | Four channel sound reproduction system |
US3761628A (en) * | 1972-04-13 | 1973-09-25 | Columbia Broadcasting Syst Inc | Stereo-quadraphonic matrix system with matrix or discrete sound reproduction capability |
US4048654A (en) * | 1976-02-18 | 1977-09-13 | Telesonics, Inc. | Stereophonic television sound transmission system |
US4139866A (en) * | 1976-02-18 | 1979-02-13 | Telesonics, Inc. | Stereophonic television sound transmission system |
US4405944A (en) * | 1980-10-14 | 1983-09-20 | Zenith Radio Corporation | TV Sound transmission system |
US4425665A (en) * | 1981-09-24 | 1984-01-10 | Advanced Micro Devices, Inc. | FSK Voiceband modem using digital filters |
US4539526A (en) * | 1983-01-31 | 1985-09-03 | Dbx, Inc. | Adaptive signal weighting system |
US4602380A (en) * | 1985-01-04 | 1986-07-22 | Cbs Inc. | Compatible transmission techniques for FM stereophonic radio and television |
US4712240A (en) * | 1986-03-26 | 1987-12-08 | Trw Inc. | Transmission and reception of television broadcast in high-fidelity stereo audio |
US4747140A (en) * | 1986-12-24 | 1988-05-24 | Rca Corporation | Low distortion filters for separating frequency or phase modulated signals from composite signals |
US4757467A (en) * | 1986-05-15 | 1988-07-12 | Rca Licensing Corporation | Apparatus for estimating the square root of digital samples |
US4760602A (en) * | 1986-10-21 | 1988-07-26 | Rca Licensing Corporation | Variable preemphasis/deemphasis network |
US4803727A (en) * | 1986-11-24 | 1989-02-07 | British Telecommunications Public Limited Company | Transmission system |
US4809274A (en) * | 1986-09-19 | 1989-02-28 | M/A-Com Government Systems, Inc. | Digital audio companding and error conditioning |
US4823298A (en) * | 1987-05-11 | 1989-04-18 | Rca Licensing Corporation | Circuitry for approximating the control signal for a BTSC spectral expander |
US4947361A (en) * | 1988-09-28 | 1990-08-07 | Unisys Corporation | Narrowband parameter estimator |
US5023609A (en) * | 1988-06-07 | 1991-06-11 | Deutsche Itt Industries Gmbh | Digital deemphasis circuit |
US5054070A (en) * | 1990-03-05 | 1991-10-01 | Qei Corporation | Stereo signal communication system and method |
US5239543A (en) * | 1990-11-05 | 1993-08-24 | U.S. Philips Corporation | Communication system and a central processing unit as well as a communication station in the communication system |
US5278909A (en) * | 1992-06-08 | 1994-01-11 | International Business Machines Corporation | System and method for stereo digital audio compression with co-channel steering |
US5282019A (en) * | 1988-10-03 | 1994-01-25 | Carlo Basile | Method and apparatus for the transmission and reception of a multicarrier digital television signal |
US5301255A (en) * | 1990-11-09 | 1994-04-05 | Matsushita Electric Industrial Co., Ltd. | Audio signal subband encoder |
US5307302A (en) * | 1991-06-03 | 1994-04-26 | Matsushita Electric Industrial Co., Ltd. | Square root operation device |
US5337196A (en) * | 1991-01-31 | 1994-08-09 | Samsung Electronics Co., Ltd. | Stereo/multivoice recording and reproducing video tape recorder including a decoder developing a switch control signal |
US5341321A (en) * | 1993-05-05 | 1994-08-23 | Hewlett-Packard Company | Floating point arithmetic unit using modified Newton-Raphson technique for division and square root |
US5357284A (en) * | 1990-03-29 | 1994-10-18 | Dolby Laboratories Licensing Corporation | Compatible digital audio for NTSC television |
US5390213A (en) * | 1990-06-01 | 1995-02-14 | Thomson Consumer Electronics, Inc. | Digital FM synthesizer for record circuitry |
US5610985A (en) * | 1993-01-22 | 1997-03-11 | U.S. Philips Corporation | Digital 3-channel transmission of left and right stereo signals and a center signal |
US5631968A (en) * | 1995-06-06 | 1997-05-20 | Analog Devices, Inc. | Signal conditioning circuit for compressing audio signals |
US5633936A (en) * | 1995-01-09 | 1997-05-27 | Texas Instruments Incorporated | Method and apparatus for detecting a near-end speech signal |
US5682431A (en) * | 1993-12-07 | 1997-10-28 | Hitachi Denshi Kabushiki Kaisha | FM stereo broadcasting apparatus and method |
US5732106A (en) * | 1995-06-05 | 1998-03-24 | Itt Corporation | Pulse-shaping filter for modulator monolithic integration |
US5796842A (en) * | 1996-06-07 | 1998-08-18 | That Corporation | BTSC encoder |
US5873065A (en) * | 1993-12-07 | 1999-02-16 | Sony Corporation | Two-stage compression and expansion of coupling processed multi-channel sound signals for transmission and recording |
US5909460A (en) * | 1995-12-07 | 1999-06-01 | Ericsson, Inc. | Efficient apparatus for simultaneous modulation and digital beamforming for an antenna array |
US6006108A (en) * | 1996-01-31 | 1999-12-21 | Qualcomm Incorporated | Digital audio processing in a dual-mode telephone |
US6037993A (en) * | 1997-03-17 | 2000-03-14 | Antec Corporation | Digital BTSC compander system |
US6259482B1 (en) * | 1998-03-11 | 2001-07-10 | Matthew F. Easley | Digital BTSC compander system |
US6434246B1 (en) * | 1995-10-10 | 2002-08-13 | Gn Resound As | Apparatus and methods for combining audio compression and feedback cancellation in a hearing aid |
Family Cites Families (12)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US3218393A (en) | 1960-02-11 | 1965-11-16 | Leonard R Kahn | Compatible stereophonic transmission and reception systems, and methods and components characterizing same |
US4539525A (en) * | 1982-09-24 | 1985-09-03 | Neil Brown Instrument Systems, Inc. | Band-pass amplifier filters |
US4799260A (en) * | 1985-03-07 | 1989-01-17 | Dolby Laboratories Licensing Corporation | Variable matrix decoder |
US4956862A (en) * | 1989-02-02 | 1990-09-11 | General Instrument Corporation | Method of providing sound privacy system compatible with mono and btsc stereo audio |
JP2992324B2 (en) | 1990-10-26 | 1999-12-20 | 株式会社リコー | Voice section detection method |
US5291289A (en) | 1990-11-16 | 1994-03-01 | North American Philips Corporation | Method and apparatus for transmission and reception of a digital television signal using multicarrier modulation |
EP0810599B1 (en) | 1991-05-29 | 2003-11-26 | Pacific Microsonics, Inc. | Improvements in signal encode/decode systems |
JPH05218992A (en) | 1992-02-06 | 1993-08-27 | Nippon Hoso Kyokai <Nhk> | Digital fm stereo modulation method |
US5373562A (en) * | 1992-08-28 | 1994-12-13 | Thomson Consumer Electronics, Inc. | Signal processor for sterophonic signals |
US5377272A (en) * | 1992-08-28 | 1994-12-27 | Thomson Consumer Electronics, Inc. | Switched signal processing circuit |
DE4417993A1 (en) * | 1994-05-21 | 1995-12-07 | Bosch Gmbh Robert | Method for transmitting digital audio signals and device for transmitting digital audio signals |
US5684881A (en) * | 1994-05-23 | 1997-11-04 | Matsushita Electric Industrial Co., Ltd. | Sound field and sound image control apparatus and method |
-
1996
- 1996-06-07 US US08/661,412 patent/US5796842A/en not_active Expired - Lifetime
-
1997
- 1997-06-02 BR BRPI9715353-2A patent/BR9715353B1/en not_active IP Right Cessation
- 1997-06-02 CA CA002255925A patent/CA2255925C/en not_active Expired - Fee Related
- 1997-06-02 BR BRPI9715316-8A patent/BR9715316B1/en not_active IP Right Cessation
- 1997-06-02 BR BRPI9714304-9A patent/BR9714304B1/en not_active IP Right Cessation
- 1997-06-02 EP EP97927918A patent/EP0974211A4/en not_active Withdrawn
- 1997-06-02 EP EP10186317.3A patent/EP2339766A3/en not_active Withdrawn
- 1997-06-02 CN CN2007101928722A patent/CN101232334B/en not_active Expired - Lifetime
- 1997-06-02 JP JP10500762A patent/JP2000513888A/en active Pending
- 1997-06-02 WO PCT/US1997/009493 patent/WO1997047102A1/en active IP Right Grant
- 1997-06-02 CN CN201210251375.6A patent/CN102890932B/en not_active Expired - Lifetime
- 1997-06-02 AU AU32262/97A patent/AU739719B2/en not_active Ceased
- 1997-06-02 BR BRPI9715315-0A patent/BR9715315B1/en not_active IP Right Cessation
- 1997-06-02 CN CNB971953422A patent/CN100362777C/en not_active Expired - Lifetime
- 1997-06-05 TW TW086107722A patent/TW401713B/en not_active IP Right Cessation
- 1997-06-06 AR ARP970102480A patent/AR013578A1/en unknown
-
1998
- 1998-03-12 US US09/041,244 patent/US6118879A/en not_active Expired - Lifetime
-
1999
- 1999-11-01 HK HK99104904A patent/HK1020240A1/en not_active IP Right Cessation
-
2000
- 2000-06-22 AR ARP000103119A patent/AR024581A2/en active IP Right Grant
- 2000-06-22 AR ARP000103120A patent/AR024439A2/en active IP Right Grant
- 2000-08-14 US US09/638,245 patent/US20080137871A1/en not_active Abandoned
-
2007
- 2007-10-30 US US11/927,734 patent/US20080095381A1/en not_active Abandoned
- 2007-10-30 US US11/927,751 patent/US8284954B2/en not_active Expired - Fee Related
- 2007-10-30 US US11/927,742 patent/US20080095377A1/en not_active Abandoned
- 2007-10-30 US US11/927,746 patent/US20080095378A1/en not_active Abandoned
- 2007-10-30 US US11/927,739 patent/US20080095376A1/en not_active Abandoned
- 2007-10-31 US US11/930,241 patent/US20080095380A1/en not_active Abandoned
-
2008
- 2008-05-14 JP JP2008127499A patent/JP4746647B2/en not_active Expired - Lifetime
- 2008-05-14 JP JP2008127502A patent/JP2008242477A/en active Pending
- 2008-05-14 JP JP2008127500A patent/JP2008242476A/en active Pending
- 2008-05-14 JP JP2008127501A patent/JP2008203891A/en active Pending
-
2009
- 2009-01-22 HK HK09100696.8A patent/HK1122660A1/en not_active IP Right Cessation
-
2011
- 2011-01-21 US US13/011,396 patent/US20110103466A1/en not_active Abandoned
- 2011-02-22 US US13/032,528 patent/US20110134992A1/en not_active Abandoned
- 2011-05-16 US US13/108,076 patent/US20110205429A1/en not_active Abandoned
- 2011-06-16 US US13/161,834 patent/US20110235705A1/en not_active Abandoned
- 2011-06-29 US US13/172,116 patent/US20110243333A1/en not_active Abandoned
- 2011-12-13 US US13/324,566 patent/US20120082206A1/en not_active Abandoned
- 2011-12-19 US US13/329,516 patent/US20120087502A1/en not_active Abandoned
-
2012
- 2012-09-07 JP JP2012197354A patent/JP2013015855A/en active Pending
- 2012-09-07 JP JP2012197355A patent/JP5538501B2/en not_active Expired - Lifetime
-
2013
- 2013-05-02 HK HK13105307.2A patent/HK1178666A1/en unknown
Patent Citations (41)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US3757047A (en) * | 1970-05-21 | 1973-09-04 | Sansui Electric Co | Four channel sound reproduction system |
US3761628A (en) * | 1972-04-13 | 1973-09-25 | Columbia Broadcasting Syst Inc | Stereo-quadraphonic matrix system with matrix or discrete sound reproduction capability |
US4048654A (en) * | 1976-02-18 | 1977-09-13 | Telesonics, Inc. | Stereophonic television sound transmission system |
US4139866A (en) * | 1976-02-18 | 1979-02-13 | Telesonics, Inc. | Stereophonic television sound transmission system |
US4048654B1 (en) * | 1976-02-18 | 1987-11-10 | ||
US4405944A (en) * | 1980-10-14 | 1983-09-20 | Zenith Radio Corporation | TV Sound transmission system |
US4425665A (en) * | 1981-09-24 | 1984-01-10 | Advanced Micro Devices, Inc. | FSK Voiceband modem using digital filters |
US4539526A (en) * | 1983-01-31 | 1985-09-03 | Dbx, Inc. | Adaptive signal weighting system |
US4602380A (en) * | 1985-01-04 | 1986-07-22 | Cbs Inc. | Compatible transmission techniques for FM stereophonic radio and television |
US4712240A (en) * | 1986-03-26 | 1987-12-08 | Trw Inc. | Transmission and reception of television broadcast in high-fidelity stereo audio |
US4757467A (en) * | 1986-05-15 | 1988-07-12 | Rca Licensing Corporation | Apparatus for estimating the square root of digital samples |
US4809274A (en) * | 1986-09-19 | 1989-02-28 | M/A-Com Government Systems, Inc. | Digital audio companding and error conditioning |
US4760602A (en) * | 1986-10-21 | 1988-07-26 | Rca Licensing Corporation | Variable preemphasis/deemphasis network |
US4803727A (en) * | 1986-11-24 | 1989-02-07 | British Telecommunications Public Limited Company | Transmission system |
US4747140A (en) * | 1986-12-24 | 1988-05-24 | Rca Corporation | Low distortion filters for separating frequency or phase modulated signals from composite signals |
US4823298A (en) * | 1987-05-11 | 1989-04-18 | Rca Licensing Corporation | Circuitry for approximating the control signal for a BTSC spectral expander |
US5023609A (en) * | 1988-06-07 | 1991-06-11 | Deutsche Itt Industries Gmbh | Digital deemphasis circuit |
US4947361A (en) * | 1988-09-28 | 1990-08-07 | Unisys Corporation | Narrowband parameter estimator |
US5282019A (en) * | 1988-10-03 | 1994-01-25 | Carlo Basile | Method and apparatus for the transmission and reception of a multicarrier digital television signal |
US5054070A (en) * | 1990-03-05 | 1991-10-01 | Qei Corporation | Stereo signal communication system and method |
US5357284A (en) * | 1990-03-29 | 1994-10-18 | Dolby Laboratories Licensing Corporation | Compatible digital audio for NTSC television |
US5390213A (en) * | 1990-06-01 | 1995-02-14 | Thomson Consumer Electronics, Inc. | Digital FM synthesizer for record circuitry |
US5239543A (en) * | 1990-11-05 | 1993-08-24 | U.S. Philips Corporation | Communication system and a central processing unit as well as a communication station in the communication system |
US5301255A (en) * | 1990-11-09 | 1994-04-05 | Matsushita Electric Industrial Co., Ltd. | Audio signal subband encoder |
US5337196A (en) * | 1991-01-31 | 1994-08-09 | Samsung Electronics Co., Ltd. | Stereo/multivoice recording and reproducing video tape recorder including a decoder developing a switch control signal |
US5307302A (en) * | 1991-06-03 | 1994-04-26 | Matsushita Electric Industrial Co., Ltd. | Square root operation device |
US5278909A (en) * | 1992-06-08 | 1994-01-11 | International Business Machines Corporation | System and method for stereo digital audio compression with co-channel steering |
US5610985A (en) * | 1993-01-22 | 1997-03-11 | U.S. Philips Corporation | Digital 3-channel transmission of left and right stereo signals and a center signal |
US5341321A (en) * | 1993-05-05 | 1994-08-23 | Hewlett-Packard Company | Floating point arithmetic unit using modified Newton-Raphson technique for division and square root |
US5682431A (en) * | 1993-12-07 | 1997-10-28 | Hitachi Denshi Kabushiki Kaisha | FM stereo broadcasting apparatus and method |
US5873065A (en) * | 1993-12-07 | 1999-02-16 | Sony Corporation | Two-stage compression and expansion of coupling processed multi-channel sound signals for transmission and recording |
US5633936A (en) * | 1995-01-09 | 1997-05-27 | Texas Instruments Incorporated | Method and apparatus for detecting a near-end speech signal |
US5732106A (en) * | 1995-06-05 | 1998-03-24 | Itt Corporation | Pulse-shaping filter for modulator monolithic integration |
US5631968A (en) * | 1995-06-06 | 1997-05-20 | Analog Devices, Inc. | Signal conditioning circuit for compressing audio signals |
US6434246B1 (en) * | 1995-10-10 | 2002-08-13 | Gn Resound As | Apparatus and methods for combining audio compression and feedback cancellation in a hearing aid |
US5909460A (en) * | 1995-12-07 | 1999-06-01 | Ericsson, Inc. | Efficient apparatus for simultaneous modulation and digital beamforming for an antenna array |
US6006108A (en) * | 1996-01-31 | 1999-12-21 | Qualcomm Incorporated | Digital audio processing in a dual-mode telephone |
US6118879A (en) * | 1996-06-07 | 2000-09-12 | That Corporation | BTSC encoder |
US5796842A (en) * | 1996-06-07 | 1998-08-18 | That Corporation | BTSC encoder |
US6037993A (en) * | 1997-03-17 | 2000-03-14 | Antec Corporation | Digital BTSC compander system |
US6259482B1 (en) * | 1998-03-11 | 2001-07-10 | Matthew F. Easley | Digital BTSC compander system |
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