US3560983A - Omnidirectional loop antenna - Google Patents

Abstract

A SINGLE TURN LOOP ANTENNA HAVING A CIRCUMFERENCE OF LESS THAN ONE-HALF OF THE WAVELENGTH OF THE ELECTROMAGNETIC ENERGY TO BE TRANSMITTED OR RECEIVED, AND AN AXIAL DIMENSION FROM 1/2 TO 1/100 OF ITS CIRCUMFERENCE.

Description

Fbiz; 7

Filed Sept. 12, 1967 E. N. WILLIE l'rrAL 3,560,983

OMNIDIREGTIONAL LOOP ANTENNA 5 Sheets-Sheet 3 FIGS.

Feb.2,1971 E. N. wmus ETAL 8,560,983"

OMNIDIRECTIONAL LOOP ANTENNA Filed Sept. 12, 1967 5 Sheets-Sheet 3 United States Patent US. Cl. 343-744 8 Claims ABSTRACT OF THE DISCLOSURE A single turn loop antenna having a circumference of less than one-half of the wavelength of the electromagnetic energy to be transmitted or received, and an axial dimension from /2 to M of its circumference.

In the prior art, single turn loops have rarely been used for purposes of omnidirectional signal transmission or reception because it was considered essential that such loops be self-resonant in order to provide the desired omnidirectionality. Because the circumference of a selfresonant loop must be at least as large as one half of the Wavelength of the electromagnetic energy to be transmitted or received, such loops are found to be physically as large as conventional linear monopole or dipole antennas while comparing unfavorably with such conventional linear antennas in power availability. In addition, the selfresonant characteristics of such loops would confine them to single-purpose or narrow band communications.

It is therefore an object of this invention to provide an omnidirectional loop antenna which is capable of handling wide band communications.

It is also an object of this invention to provide a physically compact loop antenna which may be readily incorporated into the housing of a TV, PM, or VHF-UHF communications receiver.

It is a further object of this invention to provide a compact omnidirectional loop antenna which is capable of equalling or surpassing the performance of conventional linear monopole or dipole antennas.

In accordance with the above objects the present invention provides an omnidirectional loop antenna which is formed by a single turn of electrically conductive sheet material, the circumference of the loop being substantially less than one half of the wavelengths of the electromagnetic energy to be transmitted or received and the axial dimension of the loop being from /2 to A of its circumference.

An advantage of the onmidirectional loop antenna of the present invention is in the reception of horizontally polarized signals in the VHF and UHF frequency range, such as television signals, frequency modulated broadcast signals, and other communication signals. Such horizontally polarized signals ordinarily require proper directional or angular positioning of the receiving antenna such as, for example, the commonly used dipole or monopole antennas, in order to optimize receiver performance. The antenna of the present invention obviates the need for mechanical positioning.

Another advantage of the antenna of the preset invention is its physical sturdiness as compared with conventional monopole or dipole antennas. This attribute is of particular value when the antenna is exposed to the elements as for example on a rooftop.

Because of its physical dimensions, and particularly because of the fact that its circumference is significantly less than one half wavelength, the radiation resistance and radiation reactance of the loop antenna of the present invention are far below the values which are normal in ice conventional antennas. Consequently the present loop antenna cannot be directly matched to ordinary transmission lines without serious loss of eificiency. But, by matching the present loop antenna with suitable coupling and tuning devices, normal antenna performance can be equalled or even exceeded. For example, if the loop antenna of the present invention is to be operated remotely through a trasmission line, a suitable transformer can be used to match the loop impedance to the transmission line impedance. On the other hand, if the loop antenna of the present invention is to be incorporated within the housing of a transmitter or receiver where it may be locally tuned by a suitable reactive component, the low radiation resistance of the loop makes possible a high loop Q and high resonant gain.

In either case, whether it is operated in an untuned or a tuned manner, the loop antenna of the present invention is electromagnetically conspicuous which is to say that it disturbs the local energy distribution of the E and H fields of a passing electromagnetic wave. In the untuned case, the present loop antenna acts essentially as a nonresonant, conductive ring which collects a heavy E- field flux while repelling the H-field flux. In the tuned case, the present loop antenna deflects the E field flux and collects the H-field flux. In either case, the normal E-field/ H-field energy balance of the passing electromagnetic wave is disturbed, as manifested by strong near-fields around the loop. Such disturbance causes the physically small loop of the present invention to be linked with a large aperture or effective area of interception of electromag netic energy thus providing a larger amount of available energy than an electromagnetically inconspicuous antenna such as a self-resonant quarter wave monopole or selfresonant half wave dipole of the same overall dimensions would provide.

The above and other objects and advantages of the loop antenna of the present invention will be apparent to those skilled in the art from the following detailed description and accompanying drawings which set forth the principles of the invention and, by way of example, the preferred embodiment and several modifications of the present omnidirectional loop antenna.

In the drawings:

FIG. 1 is a perspective view of the conventional Alford loop antenna of the prior art.

FIG. 2 is a perspective view of a self-resonant halfwave loop antenna of the prior art.

FIG. 3 is a perspective view of the loop antenna of the present invention.

FIG. 4 is an exploded perspective view of a transformer suitable for use with the loop antenna of the present lnvention.

FIGS. 5-7 are schematic diagrams of loop antennas illustrating the concepts of radiation inductance and radiation capacitance.

FIGS. 8-10 are perspective views of modified forms of loop antennas according to the present invention.

FIG. 11 shows a plan view and a side view of a modified form of loop antenna according to the present invention.

FIGS. 12-16 are perspective views of the loop antenna of the present invention with different tuning circuits shown in schematic form. 7

FIG. 17 is a perspective view of a further modified loop antenna according to the present invention.

FIG. 18 is a perspective view of yet another modified form of loop antenna according to the present invention.

FIG. 1 shows a conventional Alford loop antenna including two electrically parallel half-loops 1 and 2, comprising a total of four sections 3-6, each of which is approximately a quarter wave length long. The four sections 3-6 act as half the number of arms required by four resonant dipoles. The four missing companion arms are represented by four delay lines or transmission lines 7-10. Two of these transmission lines 7 and 8 are inserted in the loop at two diagonally opposed corners. The other two transmission lines 9 and 10, are part of conductors 11 and 12 which connect the two half-loops 1 and 2 in parallel to a transmission line 13 which connected the antenna to a transmitter or receiver (not shown).

FIG. 2 shows another, less common, prior art loop antenna 16 formed from a continuous wire or tubular conductor bent into a square shape and terminating at points 17 and 18 for connection to a transmission line 19. Selfresonance is established by making the circumference of the loop equal to a half wave length of the electromagnetic energy to be transmitted or received. This causes voltages and currents at terminals 17 and 18 to be in phase opposition which is suitable for connection to conductors 19a and 19b of transmission line 19. The radiation resistance of the loop antenna 16 is approximately ohms which poses a serious impedance matching problem because the practically established line impedance range limits are approximately 50 to 300 ohms. In addition, the loop 16 is highly inductive, its reactive impedance exceeding the practical 300-ohm upper limit of available line impedance. Hence loop 16 of FIG. 2 cannot be properly matched to ordinary transmission lines and, because of its line-mismatch and inductive characteristics, it cannot operate with essentially uniform in-phase currents, and is thus less omnidirectional than the Alford loop of FIG. 1.

FIG. 3 is a perspective view of an omnidirectional loop antenna 20 according to the present invention. The circumference 23 of the loop 20 is substantially less than one half the wavelength of the electromagnetic energy to be transmitted or received, and consequently the loop 20 is not self-resonant. It has been found that by substantially reducing the circumference of the loop, the resistive and reactive components of the loop impedance, are reduced in magnitude, to a point where impedance matching through a loop-to-line transformer becomes practically possible, and the reactive components of the loop impedance can be neutralized, thus making the loop an essentially resistive signal source which can be matched to the characteristic impedance of a transmission line. The impedance reduction can be enhanced by using a wide tape or sheet as the loop conductor rather than a wire or tubular conductor. The resulting increased axial dimension of the loop significantly increases its electromagnetic conspicuousness and its aperture.

The loop antenna of the present invention can then be operated in the untuned or the tuned mode, the former being preferable if a transmission line is required, the latter if the loop is an integral part of the receiver. FIG. 3 illustrates the untuned mode of operation in which nonself-resonant, low-impedance loop antenna 20 is operated in conjunction with a conventional transmission line 21, using a special impedance step-up transformer 22 to match the low loop impedance to the transmission line impedance. In FIG. 3, transformer 22 includes primary 30, which is connected to the loop terminals 31 and 32, and secondary 33, which is connected to the transmission line 21. The structural details of the preferred form of transformer 22 are set forth in FIG. 4.

FIG. 4 shows an impedance step-up transformer suitable for use in the VHF-UHF frequency range. Conventional ferrite core transformers are generally not suitable for use at such high frequencies because ferrite core material having a sufficiently high permeability but with acceptably low losses at frequencies beyond -50 mc. is not generally available in the present state of the art. Conventional air core transformers are subject to the problem of distributed inter-winding capacitances which reduce coupling efiiciency at high frequencies. This problem is aggravated, and the further problem of resistance losses is raised in the case of high impedance sources which require transformers having large numbers of turns.

The impedance step-up transformer shown in FIG. 4 avoids the problems of high frequency operation of conventional ferrite core and air core transformers. Transformer 40 includes primary winding 41 and secondary winding 42 shown in exploded form for purposes of illustration. In fact, primary 41 fits closely within opening 44 in secondary winding 42 so that both windings are concentrically disposed about axis 43. As shown in FIG. 4, the conductors employed in both windings 41 and 42 are flat tapes or sheets of electrically conductive material rather than the ordinary wires employed in conventional transformer windings. Further, the electrically conductive tape is spirally wound on itself rather than helically wound as in conventional transformers. The spiral form of the windings 41 and 42, together with their concentric relation, provides inherently good magnetic coupling between the two windings. The large surface area provided by the electrically conductive tape reduces current densities and thus keeps resistive losses low and transformer efficiency high.

The width of the conductive tape may be selected in order to achieve optimum performance at a particular operating frequency. In general, it has been found that wider conductive tapes should be used at higher frequencies, and narrower conductive tapes should be used at lower frequencies in order to achieve optimum performance. More particularly the width of the conductive tape in inches should preferably be in the range from 10 in. to 200 in? times the reciprocal of the wavelength in inches. The transformer of FIG. 4 has been successfully operated at various frequencies with conductive tapes ranging from A in. to 5 in. in width.

In general, the number of turns in the windings of a high frequency transformer should be kept to a minimum in order to minimize interwinding capacitances and also reduce resistive losses. Because of the low source impedance provided by the loop antenna of the present invention, primary winding 41 and secondary winding 42 of transformer 40 shown in FIG. 4 have lower numbers of turns than would be required in the case of a high impedance source. More particularly, it has been found that the primary winding 41 of transformer 40 should preferably have from 1 to 10 turns. Further, the turns ratio between the windings of the transformer of FIG. 4 should preferably be in the range from 1 to 10 either in the stepup or step-down sense.

It will be appreciated that, according to common practice, the individual turns of the two windings 41 and 42 may be separated by a layer or coating of electrically insulative material, not shown. It will also be apparent that, although the windings 41 and 42 shown in FIG. 4 are of rectangular cross section, windings of other cross sections such as for example, of circular cross section, may be employed. It will be further apparent to those skilled in the art that, although inner winding 41 is designated as the primary winding and outer winding 42 is designated as the secondary winding of transformer 40 shown in FIG. 4, inner winding 41 may be the secondary winding and outer winding 42 may be the primary winding within the spirit and scope of the invention.

The impedance conversion ratio of transformer 40 of FIG. 4 is where 11 and n are primary and secondary turn numbers and K the voltage or current coupling factor between the windings. With a 5:2 turn ratio, as shown in FIG. 4 and K an effective impedance conversion of approximately 5:1 can be obtained, such as 10 ohms loop impedance into 50 ohms transmission line impedance. In practical experimentation between 60 and 220 me. have exceeded this moderate impedance conversion ratio by far, using ratios of actual turns such as 14:2, 16:1 etc.

Practical experience has also shown that the width of the conductor tape in the transformer should be a reasonable fraction of the width of the conductor tape of the antenna loop. For example the width of the transformer conductor should be from 3% to 100% of the width of the loop antenna conductor and preferably -20% of the loop conductor width.

Before describing the tuned mode of operation of the present non-self-resonant, low impedance loop, it will be useful to give a brief analysis of the various linkages of the loop as a whole and its individual sections with the H and E-fields of a passing electromagnetic wave. Referring to FIG. 5, the H-field linkage with the loop antenna 51 develops an electromotive force e between its terminals 52 and 53 by being linked with an H-field and the H-flux which it encompasses. The H-flux is perpendicular to the plane of the loop and is represented in FIG. 5 by a second, inner loop 54 connected to a magnetic inductor H which is linked with the antenna 51 by transformer action. If loop antenna 51 is small in terms of a wavelength, and if the center of the flux happens to be inside the loop 51, the induced e is equal to the sum of the four equal, in-phase E.'M.F.s, e -e which are induced in the four sections 56-58 of loop antenna 51. FIG. 6 shows a loop antenna 51 wherein the secondaries 61-64 of the four transformers 65-68 are inserted into the four loop sections 56-58. The primaries 69-72 of transformers 65-68 are connected to four magnetic inductors H -H which are assumed to be identical and in phase. The induced E.M.F.s, e -e are shown across transformer secondaries 61-64.

In FIG. 5, transformer action between the antenna 51 and the loop 54 represents the total radiation inductance of loop 51, i.e., its mutual inductance with inductor H in space. In FIG. 6, the transformer action between secondaries 61-64 and primaries 69-72 represent the lumped radiation inductance of the individual loop sections 55- 58, i.e., their mutual inductances with inductors H -H in,

space. It should be noted that the four inductors H -H are floating. No reference to earth is indicated.

Referring to FIG. 7, the E-field linkage with the loop antenna 51 is shown in schematic form. Each of the sec tions 55-58 of loop 51 has radiation capacity, Le, a linkage with the E-field through an equivalent, lumped capacitor C -C The E-field, as a voltage source, is represented by four voltage generators e -e each of which is referred to ground. If the voltages are of equal magnitude and in phase, and if travel time along the circumference of the loop 51 is negligibly small with reference to a wavelength, the capacitively induced voltage e between the terminals 52 and 53 of loop 51 will be negligibly small, or zero. On the other hand, if the dimensions of loop 51 are large enough to create phase differences between the four generators (a -e differential voltages will be developed between them and currents will flow im the various loop sections 55-58. A voltage will thus be developed across the impedance of each section 55-58, and a resulting capacitively induced voltage e will be observed between terminals 52 and 53 of loop 51. If delayline action within the loop sections 55-58 is ignored, this voltage would be e =2Ke sin assuming that e =e =e =e that is the phase shift between opposite generators, and K is a constant determined by the impedance of the sections 55-58.

The participation of individual generators e -e in the formation of output voltage e is governed by the direction of travel of the received wave. If the direction of travel is as indicated by arrow W generators e and 6E3 deliver the differential signal voltage, but if the direction of travel is W; the other two generators e and e develop the signal voltage.

The radiation capacities C -C of the sections 55-58 of loop 51 play a dual role. Because of their linkage with the E-field, radiation capacities C -C act as signal sources. At the same time they act as elements of the delay lines which interconnect these signal sources. The distributed radiation inductance and distributed radiation capacity of the loop are the reactive elements necessary to make an effective near-loss-free delay line. The groundreferences of the radiation capacities furnish the necessary reference terminals for the delay lines.

Since the passing electromagnetic wave induces out-ofphase voltages which are connected in parallel by delay lines, local standing waves are set up within the loop which tend to increase the observed capacitively induced output voltage. There are many complicated aspects of this phenomenon including the occurrence of simultaneous travel of delayed signal voltages and currents in opposite directions over the same delay lines section, from the leading capacitive signal source to the lagging source, and from the lagging source to the leading one. The beneficial effect of the loop delays may be further enhanced by wafiling the conductor of the loop in order to increase its inductance, or by adding capacitive targets to the lOOp in order to increase the delays. Such modifications will be described in greater detail hereinafter.

Referring again to FIG. 6, in which the four transformers 65-68, represent the four radiation inductances which link the sections 55-58 of the loop 51 with the H-field represented by the four equivalent iniductors H -H it can be seen that the four induced voltages e -e will not be equal at a given moment, nor in phase, if loop dimensions are sufficiently large to create observable phase shifts between these inductors. For instance, if the direction of wave travel is W inductor H will lead inductor H and if all delay line effects are ignored, the output voltage would be E ZC Sin /2 assuming that e e while e and e do not contribute because of the direction of travel of the wave.

Interconnection of the two inductors e and e and their associated loop sections 55 and 57 is in series, not in parallel as it is in the case of the E-field linkage. Delay lines again modify the actually obtained voltage through creation of standing waves. In this case the two induced voltages e and e are connected in subtractive rather than additive relation, and this relation is modified by the simultaneous travel over the same delay lines of the leading voltage or current towards the lagging source, and the lagging source with further delay towards the leading source.

Assuming that the simplified magnetic linkage of the loop with a homogeneous H-field shown in FIG. 5 is the sole source of energy, the available power P of the loop antenna of the present invention is given by:

6H2 P =1 n. A Rt where R is the radiation resistance of the loop. e being proportional to the flux-intercepting area of the loop changes by the square of loop circumference. Therefore, it is apparent that if radiation resistance remains constant with changing loop circumference L available power would decline by the fourth power of loop circumference, or:

where K is a constant. In reality the radiation resistance R is not constant. Instead, it decreases by approximately the fourth power as the loop circumference decreases below a half-wave-length making the available power P,, approximately constant. The radiation resistance of the present loop antenna is approximately 12 ohms at a circumference of a half-wavelength and declines by the fourth power of loop circumference. Radiation resistances of single ohm and even fractional ohm values are therefore commonly encountered in the loop antenna of the present invention.

The reactive characteristics of the present loop antenna are usually inductive in nature and depend upon loop circumference and conductor width. An increase of conductor width decreases inductance and increases radiation capacitance, thus tending to neutralize the inductance. Complete neutralization is, however, not achieved until conductor width approaches one-half of loop circumference as in the case of the loop antenna 81 shown in FIG. 8. The loop antenna 81 of FIG. 8 is essentially nonreactive and can be matched to any conventional transmission line 82 by a transformer 83, preferably of the type shown in FIG. 4. As indicated, circumference C of the loop 81 is less than a half wave length, while height H is half of C or twice the distance D between parallel walls. The nonreactive properties of loop 81 over a wide frequency range are similar to the properties of a typical single wire transmission line. The unit-length inductance and unitlength radiation capacity of the loop 81 act as incremental reactive parameters and its distributed radiation capacity acts as a common reference terminal. The characteristic impedance of the loop 81 is on the order of ohms or even fractional ohms, thus requiring transformation to the higher characteristic impedance of a line to which the loop is connected such as by means of a transformer of the type shown in FIG. 4.

High impedance step-up conversion ratios such as required by the wide loop of FIG. 8 can be avoided, and the corresponding high-current losses in interconnection between the loop-to-line matching transformer can be reduced, by reactive neutralization on a sectional basis along the circumference of the loop as in the case of loop 90 shown in FIG. 9. Loop 90 has four sections 91-94 which are interconnected at the four corners 95-98 of the loop. The loop conductor varies in width to form four large capacitive targets 99-102 which are preferably equally spaced from each other.

The capacitive targets 99-102 are from one to 10 times as wide in the direction of the axis of loop 90 as the remaining section 103-106 connecting them. Each of the capacitive targets 99-102 preferably comprises an equal portion of the circumference of loop 90. More particularly, each of the four capacitive targets 99-102 may comprise from A to of the circumference of loop 90. It will be appreciated by those skilled in the art, however, that a greater or lesser number of capacitive targets may be employed within the spirit and scope of the present invention. In general, the capacitive targets should relatively comprise from /2 to of the circumference of the loop.

The radiation capacities of capacitive targets 99-102 perform several roles simultaneously. (a) They link the loop with the E-field. (b) They reduce, or neutralize, inductances of the four narrow loop subsections 103-106, which interconnect the capacitive targets 99-102, thus giving the loop a reduced total inductance. (c) They act as elements of delay lines having greater delay than have the loops previously shown. The increased delay increases phase shifts between the capacitive linkage with the E- field and the magnetic linkage with the H-field of the passing electromagnetic wave thus resulting in larger differential signals. These may be traded off against a further reduction of loop dimensions, if loop miniaturization is desirable.

The whole loop 90 of FIG. 9 can also be described as a delay line with lumped reactive components, in this case four inductors and four capacitors. While over a very wide frequency band such a loop fails to equal the purely resistive properties of a delay line with distributed incremental reactive parameters, it is, nevertheless, an essentially purely resistive signal transmitting line over a reasonably wide frequency band. The advantage of the loop 90 of FIG. 9 over the wide loop of FIG. 8 is its higher impedance which is on the order of ohms and low tens of ohms rather than fractional ohms and low ohms, and a corresponding reduction of losses in the interconnection between the loop and the primary of the matching transformer due to the lower current densities.

FIG. 10 shows a modified loop in which sections 103- 106, interconnecting the capacitive targets 99-102, are waffied or corrugated in order to increase conductor length and inductance without increasing the circumference of the loop. Greater delays and larger differential signals are therefore developed.

The loop antenna of the present invention is preferably operated in the tuned mode if the loop can be located near the receiver or transmitter, or if it is made an integral part of such equipment. In the untuned mode of operation described above, the radiation resistance of the loop acts as the source of available signal power which must be matched to the characteristic impedance of a transmission line if maximum signal voltage or current is to be delivered to the input terminals of the first amplification siage. In the untuned mode of operation, reactive components, such as loop inductance, should be minimized or neutralized, if maximum energy transfer is to take place.

On the other hand, in the tuned condition, energy considerations assume a minor role, since the first amplification stage is normally a voltage-sensitive device such as an electronic tube or current-sensitive device such as a transistor. In the tuned condition, radiation resistance assumes a major role as an attenuator of the resonant circuit. Fortunately radiation resistance decreases very rapidly, as loop circumference is decreased. Loop inductance, on the other hand, decreases much more slowly unless extremely broad loop conductors are used. The net result is a substantial increase of loop Q, which is important if increased resonant gain of a smaller loop is to compensate for a reduction of the induced E.M.F., due to reduced loop area and consequent reduced flux linkage. Since, in theory, the flux-induced E.M.F. declines with the square of loop circumference and Q increases approximately by its third power, a linear net voltage increase should appear at the terminals of the tuned loop, in inverse proportion to its circumeference. In fact, this voltage increase is normally offset by other factors such as increased radiation capacity and losses in the loop and its interconnections due to increased current density resulting from reduced impedance, etc. However, experiments have verified that a surprisingly small, tuned loop is capable of delivering surprisingly large resonant voltages, or currents.

As mentioned in connection with FIG. 10, loop inductance can be increased without increasing overall loop dimensions by wafiling or corrugating the loop conductor.

FIGS. 11a and 11b show another wafiied loop 111 in which wafiling approximately triples the conductor length thereby increasing loop inductance between terminals 112 and 113 without increasing the circumference of the loop 111 and only slightly increasing its radiation resistance. Neither radiation capacity nor radiation inductance are appreciably changed by waffiing the loop conductor in FIG. 11, but the self-inductance of the conductor is substantially changed. Radiation inductance and radiation capacity are general contour phenomena and do not follow minor surface variations, such as produced by wafiiing. The increased self-inductance of the four sections 114- 117 of loop 111 in FIG. 11 due to wafiling increases the delay line characteristics of these sections. Because of increased self-inductances with nearly unchanged sectional radiation capacities, the phase shifts between the magnetically and electrostatically induced voltages are substantially increased thus increasing the output signal across terminals 112 and 113 of the loop.

FIG. 12 shows a tuned, sub-half-wave-circumference loop 121 which is shunted by a variable tuning capacitor 122 and connected to the differential input terminals 123 and 124 of amplifier 125, having a common input reference 126 which is connected to ground. It will be appreciated that capacitor 122 may be replaced by a selector switch and individual capacitors such as might be found in an input tuner of a television receiver.

FIG. 13 shows a similar loop 121 feeding into a single-ended-input amplifier 131. One loop terminal 132 is grounded while the other 133 is connected to the ungrounded input terminal 134 of the amplifier. Variable tuning capacitor 122 or an equivalent capacity selector switch is connected in parallel with the loop terminals 132 and 133.

In FIG. 14 an impedance-converting transformer 141 is connected between the loop terminals 132-133 and variable tuning capacitor 122. Such an arrangement may be desirable in order to avoid inconveniently large tuning capacities which would otherwise be required by the low loop inductances. Transformer 141, which may be of the type shown in FIG. 4, steps up the loop inductance by the square of the turns ratio between primary 142 and secondary 143 and thus reduces the required tuning capacity. Connection to a transmitter or receiver is indicated by arrows 144 and 145.

In FIG. tuning capacitor 122 of FIG. 14 is replaced by a fixed capacitor 151, which may be a separate component or simply an incident input shunting capacity between the input terminals of an input tube or transistor. Tuning or station selection is effected by variable inductor152, which, if desired, may be replaced by a selector switch with individual fixed inductors such as, for example, in the input tuner of a television receiver.

The circuit of FIG. 16 is similar to that of FIG. 15, except that the loop-transformer 141 of FIG. 15 is omitted.

The large conductor Width of the sub-half-wave, singleturn loop of the present invention can give rise to eddy current losses in the conductor surface. This condition is most likely to occur if the E or H-fields sweeping over the loop are not truly at right angles to the loop axis. Wave tilts, local reflectors absorbers or reradiators may create field distortions, or concentrations thereby causing nonhomogeneous current densities within the loop conductor and consequent eddy current losses.

FIG. 17 shows a loop antenna 171 divided into narrow parallel loops 172-178 which serve to reduce such eddy current losses and thus increase available energy in the case of an untuned loop, or improve Q in the case of a tuned loop. The individual, narrow loops 172-178 are connected in parallel by two bus bars 179 and 180, which are connected to the loop terminals 181 and 182. If desired, an impedance step-up transformer 183 may be used to connect loop 171 to suitable tuning circuitry or to a trans-mission line.

Although the loop antennas shown in FIGS. 3-17 have rectangular cross sections, it will be appreciated that other cross sections such as, for example, circular or elliptical cross sections may be employed within the spirit and scope of the invention. Omnidirectionality of some oblong shapes may not be perfect but may be adequate for specific purposes. The linkage of the loop antenna of the present invention with both the H and E-fields of a passing electromagnetic wave gives assurance that complete failure to receive a signal under the most adverse conditions is much less likely than in the case of conventional loops or linear antennas.

Among the various possible odd 100p shapes, falling within the scope of invention, is one which is of particular importance to aviation, space-flight and for submarine purposes. It is illustrated in FIG. 18.

Loop antenna 191 of FIG. 18 is similar to the essentially nonreactive loop antenna 81 of FIG. 8, but instead of having a rectangular cross section, loop antenna 191 of FIG. 18 has a tear-drop shaped cross section which is suitable for applications in which air-resistance or water resistance should be minimized. For example the loop antenna 191 of FIG. 18 may be made flush with the surface of an airplane wing or the periscope of a submarine 1 0 with blunt edge 192 as the leading edge and sharp edge 193 as the trailing edge.

It will be recognized by those skilled in the art that other and further modifications and adaptations of the form of the loop antenna of the present invention and its associated circuitry may be made without departing from the spirit and scope of the invention as set forth with particularity in the appended claims.

What is claimed is:

1. An antenna for transmitting or receiving electromagnetic waves comprising a sheet of electrically conductive material formed into a single continuous loop, the circumference of said loop being less than one half of the wavelength of the electromagnetic waves and the axial dimension of said loop being in the range from to of the circumference of said loop, selected sections of said loop having an axial dimension greater by a factor in the range from two to ten than the remaining sections of said loop.

2. The antenna of claim 1 wherein said selected sections cumulatively comprise from to of the circumference of said loop.

3. The antenna of claim 2 wherein said selected sections comprise equal portions of the circumference of said loop.

4. The antenna of claim 3 wherein said remaining sections separating said selected sections are of equal length.

5. The antenna of claim 4 wherein said remaining sections are wafiled to increase their length by a factor in the range from one and one half to four.

6. The antenna of claim 1 further comprising a variable capacitor coupled across the ends of said loop for tuning said antenna.

7. An antenna for transmitting or receiving electromagnetic waves comprising a sheet of electrically conductive material formed into a single continuous loop having a circumference less than one-half of the Wavelength of the electromagnetic waves and an axial dimension from to ,4 of the circumference of said loop, said loop including four selected sections having an axial dimension from two to ten times that of the remaining sections, said selected sections being equally spaced about the circumference of said loop, and each of said selected sections comprising from to of the circumference of said loop.

8. The antenna of claim 7 wherein said remaining sections are waflied to increase their length by a factor in the range from one and one-half to four.

References Cited UNITED STATES PATENTS 926,934 7/1909 DeForest 343-850 2,493,569 l/l950 Brown, Jr. 343-802X 2,501,430 3/1950 Alford 343-741 2,515,816 7/1950 Zindel, Jr. 343-866X 2,618,746 11/1952 Pauch 343-743X 2,622,196 12/1952 Alford 343-855X 3,389,395 6/1968 Lejkowski, Sr 343-743 FOREIGN PATENTS 761,557 11/1956 Great Britain 343-866 OTHER REFERENCES The A.R.R.L. Antenna Book, West Hartford, Conn., 1960, The American Radio Relay League TK6565 A6A6, title page, reverse side of title page, and pp. 65-66.

ELI LIEBERMAN, Primary Examiner N. NUSSBAUM, Assistant Examiner US. Cl. X.R. 343-866, 899

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