US3643163A

US3643163A - High-order mixer and comparator

Info

Publication number: US3643163A
Application number: US10505A
Authority: US
Inventors: George Bruck
Original assignee: Avco Corp
Current assignee: Avco Corp
Priority date: 1970-02-11
Filing date: 1970-02-11
Publication date: 1972-02-15
Anticipated expiration: 1989-02-15

Abstract

A high-order mixer uses a full-wave, odd-function, exponential (antisymmetric) nonlinearity to transfer modulation information onto a carrier or reference signal. In one form, a pair of highorder mixers are used in a balanced arrangement to effect comparison of the amplitudes of two high-frequency input signals without regard to the phase of these signals and to provide an output at the reference frequency (or a multiple) which (i) increases monotonically as the difference between the signal amplitudes increases, (ii) reverses phase as the signal amplitude difference reverses, and (iii) becomes zero when the two signal amplitudes are equal. In another example, such a comparator is used for automatic tuning.

Description

United States Patent Bruck Feb. 15,197

[54] HIGH-ORDER MIXER AND COMPARATOR George Bruck, Cincinnati, Ohio [52] US. Cl "325/177, 325/449, 332/46,

332/52 [51} Int. Cl .1104!) l/04, H03d 7/02 [58] Field ofSearch ..325/l7,105, 148,174,177,

2,874,274 2/1959 Adams et al. ..325/174 X 3,355,667 1 H1967 Bruene ..325/174 3,475,703 10/1969 Kennedy et a1, ..333/17 3,509,500 4/1970 McNair et a1. ....325/l74 X Primary Examiner-Robert L. Richardson Attorney-Darby & Darby [57] ABSTRACT A high-order mixer uses a full-wave, odd-function, exponential (antisymmetric) nonlinear-ity to transfer modulation information onto a carrier or reference signal. In one form, a pair of high-order mixers are used in a balanced arrangement to effect comparison of the amplitudes of two high-frequency input signals without regard to the phase of these signals and to provide an output at the reference frequency (or a multiple) which (1) increases monotonically as the difference between the signal amplitudes increases, (ii) reverses phase as the signal amplitude difi'erence reverses, and (iii) becomes zero when the two signal amplitudes are equal. In another example, such a comparator is used for automatic tuning.

19 Claims, 6 Drawing Figures DIO PHASE FREQUENCY AMPLIFIER P1 DETECTOR l I '1'] l! I lf54 58 57 I I l 53 55 I l II 73 5KHz REFERENCE TUN'NG FREQUENCY Q J- OSCILLATOR 74 VIE 5KHZ T PAIENIEDFEB is 1912 3,643 163 sum 1 0F 2 HSUPPRESSOR v CIRCUIT V COS WM 20 1m v COSLLI 1 2 B B 26 H CURRENT F a G. 2 VOLT VOLT INVENTOR. G EORGE B RUCK CURRENT ATTORNEYS PATENIEDFEB I 5 I972 a, 643, 1 s3 SHEET 2 [IF 2 I FIG..4

RF A TUNABLE i INPUT FILTER I 32 VOLTAGE CURRENT I /36 sENsOR sENsOR TUNING CONTROL VRA VRB SYSTEM LUC+ 90 LUC I T 34 I 33% HlGIMllgIzDER PHASE (COMPARATOR) e DETECTOR FIG. 5

l i ....I i1 P 52 I :.IZ.I[ 58 ..I..I. 57

5| I VZ/v-v T T I 1 T f L I I I I II SKHZ TUNING I SEEEBEIIE? OSCILLATOR SYSTEM 4' 5KHZ 7O AUIJIO PHASE FREQUENCY INVEN AMPLIFIER DETECTOR GEORGE 585% ATTORNEYS HIGH-ORDER MIXER AND This invention relates to mixers and, more particularly, to circuits which use high-order nonlinear elements in mixers and signal comparators.

In a number of applications such as for the detection of impedance characteristics of circuit networks and other similar apparatus, there exists the problem of comparing two signals (of the same frequency or different frequencies) in absolute magnitude without regard to their relative phase. In one such application these are radiofrequency (RF) signals.

, A simplistic approach known in the past was to rectify the two signals and compare the two outputs. This approach is not satisfactory for low-level signals because these signals cannot be properly rectified. Even for large enough input signals, drift in amplitude is still a significant problem and frequently is a limiting factor. DC amplification, while possible, generally introduces error.

One approach toward solving this dilemma is by a mixing technique. Each of the signals to be compared is mixed with a local oscillator signal. The resulting difference-frequency signals are suitably amplified in individual amplifiers, one for each signal, and the rectified outputs of these amplifiers are compared. This technique has two inherent limitations. First, the local oscillator frequency must properly track the signals being compared in order for the difference-frequency signal to be of the proper frequency. While this can be done by locking a signal proportional to the difference between the frequencies of the incoming and fixed local-oscillator signals to a second local oscillator, this is extremely complex. Second, the

two difference-frequency amplifiers must have and maintain the same gain which requires significant expense and introduces difficulty in the design and manufacture of these am plifiers.

The present invention accomplishes such a precision comparison by an extremely simple circuit utilizing high-order mixer techniques.

The present invention is also directed to a high-order mixer for cross-modulating two signals, which inherently avoids the difficulty of frequency drift of either of the signals relative to the other.

It is an object of the present invention to provide a circuit for comparing the amplitudes of two signals without regard to their relative phase. This is also applicable in many cases to signals of different frequency.

It is another object of the present invention to compare two signals of the same frequency using mixer techniques and to provide an output signal which monotonically increases with increasing difference between the amplitudes of the two signals, which reverses phase if the amplitude difference reverses in polarity and which provides a null output when the two signals are identical.

It is a further object of the present invention to provide a circuit for the comparison of two signals using mixer techniques where the output signal is approximately proportional to the difference between the squares of the amplitudes of the input signals and is at a fixed frequency independent of the two signals.

It is an additional object of the present invention to provide a circuit for cross-modulating two signals regardless of frequency drift of one of the signals.

For a better understanding of the present invention, together with other and further objects thereof, reference is made to the following description taken in connection with the accompanying drawings, and the scope of the invention will be pointed out in the appended claims.

In the drawings:

FIG. Ia represents a schematic drawing in partially block diagram form of a high-order mixer according to the present invention;

FIG. 1b represents a schematic circuit diagram of the highorder mixer in more detail;

FIG. 2 is a graph showing the current versus voltage relationship of a nonlinear means used in the present invention;

FIG. 3 is a schematic circuit diagram of a signal comparator forming one aspect of the present invention;

FIG. 4 is a block diagram of a tuning control circuit using the signal comparator of FIG. 3;

FIG. 5 is a more detailed schematic circuit diagram of the arrangement of FIG. 4.

The basic circuit of the high-order mixer is shown in FIGS. l and 2. The signal comparator mentioned above is shown in FIG. 3. Understanding of the merits of this invention will be aided by considering the mathematical derivation of the output signal of the high-order mixer. Since the same mathematics explains the derivation of the signal comparator output of FIG. 3, the output of the mixer will be first derived and the applicability of the derivation to the signal comparator will then be shown.

The high-order mixer, shown in bloclk form in FIG. la, is formed principally of a nonlinear means schematically represented at .11. Nonlinear means II preferably has a characteristic approaching that shown in FIG. 2 where the current i passing through the nonlinear means is an odd function of the voltage developed across the means. That is, i=f v =f(v). The positive quadrant of the graph of FIG. 2 (where both voltage and current are positive) illustrates a monotonically increasing exponential functional relationship between v and i. The negative quadrant (where both v and i are negative) shows a monotonically decreasing exponential functional relationship between v and i. This characteristic has double-mirror symmetry (antisymmetric), about both the vaxis and the i-axis.

Two signals are supplied to the nonlinear means 11, to be mixed by it. A first signal V is one having a predetermined frequency, and an amplitude which varies according to predetermined data or intelligence. It may for example be a modulated carrier for conveying information or a control signal for operating suitable control circuits. The second signal V is a local or reference alternating signal, corresponding to an intermediate-frequency source or local oscillator. The circuit of the invention may operate to cause the local signal (or a harmonic of it) to vary in accordance with the am plitude variations of V In the illustrative form of circuit represented in FIG. la, the signals V A and V are additive, with the respective input terminals to which they are applied being arranged in series with one another and the nonlinear means ll. Also shown illustratively in series is a DC suppression circuit 10 which permits only the AC component of current to flow through the nonlinear means 11. As shown below, a simple form of DC suppressor circuit is merely a series capacitor, although other circuits, including transformers and the like, may be used.

The output developed by the nonlinear means II is taken from its current, illustratively by an output transformer 12 whose primary is in the series circuit. Other forms of currentresponsive output may be used, such as resistors, transistors or others. However, use of a transformer is convenient, since it permits a grounded connection to its secondary which is desirable.

Referring to FIG. lb, one practical application of the block diagram of FIG. la is shown. Here the desired characteristic of nonlinear element 11 is achieved by the composite characteristics of a pair'of oppositely polarized parallel-connected semiconductor diodes. Other nonlinear elements are also feasible, such as appropriately sloped vacuum tube diodes or properly designed thyrite elements (made of a polycrystalline carborundum structure having semiconductor properties).

The two diodes, D and D have associated characteristics described by the following equations:

i, Me 1 for positive values of v; i. a O

for negative values of v. (A)

'2 M1 e-") for negative values of v; [1 E 0 for positive values of v. (B)

where v qVlkl; q is the charge of one electron, k is Boltzmans constant, and Tis the absolute temperature in degrees Kelvin, making q/kt approximately equal to the value 40 at room temperature. V, in this case, is the voltage applied across the diodes, and i is a reference current which depends upon the size of the diode junction.

The voltage V applied across the diodes normally exceeds 10.5 volt because of the usually large reference voltage V (or is made to exceed 10.5 volt). This causes the exponential term a" to be much larger than I, and the currents through the diodes then increase with voltage quite accurately according to the exponential relationship.

Then, (considering only the nonzero relationships for i, and 2),

In a practical situation, the reference currents i,, of the two diodes are rarely identical. The DC blocking capacitor C avoids any detrimental action in this regard and also removes any DC component of V,,. This can be shown by arbitrarily setting i =i e and i =i,,e' (3); (4) where A is some discrepancy factor. Then, designating the two components of the applied voltage by the subscripts AC and DC, it follows that i int? AC e' 'm i2 inr e ee-Pal: (6)

Since the blocking capacitor prevents DC from flowing m= (7) Then it follows that 2 u Thus the blocking capacitor C eliminates DC as well as overcomes most imbalance in the diodes D, and D to establish the desired ideal exponential relationships.

The composite current i through the diodes, which also flows in the external circuit by way of the transformer primary WAC I/ AC e e The applied voltage across the two parallel diodes is VAC=VA cos W t+V cos WA (l2) Substituting 12) in l l and recognizing that v =4O V v =4O V, and v,.=40 V then i =2i Sinh [4OV cosW t+4OV cosWd] (13) Using the relationship Sink (0+8) Sinh a Cosh [3+ Cosh a Sinh B (14) Expanding the exponential terms of 17) by using equation 9.6.34 on page 376 of the Handbook of Mathematical Functions, edited by Abramowitz and Stegun and published as Applied Mathematics Series 55 by the National Bureau of Standards, 7th printing. May i968, and letting Z=v and O=W1 one obtains Ml-k211i") cos air-2W cos Zwl+ .1

Similarly, by using the relationship Cosh a='/(e +e and substituting a=Zcos0 one obtains Cosh (Zcos0)=/(e +e' Expanding again by using equation 9.6.34 of Abramowitz and Stegun there results Cosh (vcos Wt )-I,,( v)+Z[ I v)cos2 W!+I,( v )cos4 Wl+.

Substituting these expansions of sinh and cosh in i, in equation 15) one obtains ill) The I,(v term cannot be simply approximated since V is greater than 0.5 volts and the series development has to be carried too far. As this is only a multiplying factor, it can be lumped together with i,, into a fixed value for a fixed applied If V is small enough, the expansion of l,,( V may be truncated after the second term. One then obtains the following approximation of the total output current:

I 2 i =4iuIl (401/ [1 COS (or! (28) r')i m .l i r where K is a constant and f( V,.) means a function of V,., which is constant.

This is the result of the analysis.

Thus, the amplitude of the input signal, including any modu lation thereon, has been transferred onto the local signal according to the relationship (29). In effect, the local signal W, is amplitude modulated by the square of the input voltage modulation V For best results a filter capacitor C should be placed across the output transformer Tto eliminate higher harmonics of W... Where W is larger than W,., this will also eliminate other un wanted frequency components. In addition, the source resistances and impedances of all capacitors should be kept small to preserve the above relationships.

Since any internal diode resistance will cause the diode characteristic to deviate from the desired exponential relationship, such resistance should also be kept to a minimum. Schottky diodes are designed for low noise applications so that their internal resistances are substantially lower than for the ordinary diode. Such diodes are therefore particularly suitable for use as the nonlinear means of "to present invention.

Referring now to NH. 3. an im mrlaul clnlmtlinu'nl ol' the high-order mixer is shown, which is in the nature of an amplitude-comparison circuit. In a comparator circuit, the absolute amplitude of each signal being compared is not materi al, and it is sufficient to provide a resultant output which merely monotonically increases with the increasing difference between the amplitudes of the two signals, reverses phase if the difference in amplitudes reverses in polarity, and properly nulls if the two signals are equal in amplitude. Such a result is derived from a signal proportional to the algebraic difference between the squares of the two signals to be compared, which is the output of the FIG. 3 circuit.

An examination of the circuit of FIG. 3, will show that this circuit is no more than two high-order mixers of FIG. lb, each having a separate input signal, with a common local signal as reference V cosW t and v cosW,,r are the two signals to be compared in amplitude without regard to their phase or frequency. The V input is connected to a series circuit including a blocking capacitor 23 and a nonlinear means having an odd-function, exponential characteristic shown as

diodes

19 and 20. The V signal is connected to the other series circuit including the blocking capacitor 26 and nonlinear

means comprising diodes

21 and 22. The two series circuits are connected to the outer terminals of a center-tapped output transformer 29. Across the half windings of the transformer primary are filter capacitors 27 and 28. A local signal source is placed effectively in series in each series circuit by transformer 25.

Using Equation (25) for i derived previously for the single mixer case, the following oppositely directed currents are produced at the circuit output due to the action of each side:

where the subscripts A and B are designations for the respective sides of the circuit.

The net output current in the secondary of transformer 29 is proportional to Using the same expansion for 1,, as previously (equation 27) and truncating the expansion after the second term, the following approximation for the output is valid:

where K is a constant and f(V,.) denotes a function of V,., which is kept fixed.

This output then is the desired signal, since it is of fixed frequency (that of the local source W,.) and proportional to the difference between the amplitudes of the applied input signals (squared), will reverse phase when the difference between the input amplitudes reverses in polarity, and will be zero when those amplitudes are equal.

Such a signal is particularly applicable as an error or command signal in a closed loop control system as will be described with regard to the system shown in FIGS. 4 and.5.

Referring to FIGS. 4 and S, a practical application of the comparison circuit shown in FIG. 3 is shown. These figures depict circuitry useful in the automatic tuning or frequency control ofa radiofrequency tuning circuit such as might be used in a radio transmitter. A similar environment in which this invention is useful is in automatic tuning apparatus such as for tuning the output of a power amplifier to a resonant and matched condition with an antenna system by the use of a resonant circuit such as a pi-type filter network similar to that shown in FIG. 5.

FIG. 4 illustrates a block diagram of a portion of the automatic tuning system. A radiofrequency input signal (such as derived from an oscillator or modulator) is supplied to an amplifier 30 whose output provides a voltage sample and a cur rent sample by operation of voltage sensor 31 and current sensor 32, respectively. The amplifier output is also coupled to a tunable filter network 37 whose output is fed to an antenna 39. The filter 37 is adjustable under the control of tuning control system 36 which in turn is controlled so that the antenna 39 will become tuned and matched to the source resistance presented by amplifier 30. The amplitude and phase relationships of the current and voltage samples are used to establish the matched and tuned condition of filter 37.

In one approach to tuning and matching a filter, it is desired that the amplitude of the vectorial sum of the voltage sample and the current sample be compared with the amplitude of the voltage sample alone. When these two values are made equal by operation of the system shown in FIG. 4, one requirement in achieving a tuned and matched condition of filter 37 is met.

The voltage sample V and a voltage proportional to the sum of the current sample V and the voltage sample V, form the inputs to the comparator 33, which incorporates the high-order mixer described above. The output of the comparator is designed to be at null when the input signals to the comparator are equal in amplitude. When these input signals are not equal, indicating that the filter has not been properly adjusted, an error signal, e, is produced at the output of the comparator. This signal is amplified by an amplifier 34 and detected by phase detector 35 to produce a. DC error signal. This resultant signal is used by the tuning control system 36 to adjust filter coils or capacitors in the filter network to bring the filter into a resonant and matched condition.

Referring now to FIG. 5, a more detailed circuit is shown which may, for example, form the final amplifier section of a radio transmitter. An RF input signal, which may be from an exciter, is applied to the control grid of a conventional amplifier tube 41 to a coupling capacitor 42. Bias, screen, and plate voltages are supplied to the tube 41 by means of respective RF chokes 43, 45 and 46. The output from amplifier tube 41 is supplied to a coupling capacitor 47 and. a coupling coil 50 to the tunable filter network 37 shown as consisting of a variable inductance 56 and selectable capacitor sets 52, 54, 58 and 53, 55, 57. The RF output of the filter network 37 may be used to drive the antenna 39 or an antenna coupler stage or other utilization circuit. It is desirable that the filter network 37 be kept tuned to the RF signal input to it and this is accomplished by automatically varying inductor coil .56 and

capacitors

52, 54, 58 and 53, 55, 57 through the use of the tuning control system 36 as described below. While a pi-type filter is shown as output filter 37 this discussion applies to any similar tuned network which may be used in this particular application. Narrow band networks from simple LC (L standing for inductance and C capacitance) to tee or pi networks discussed here are suitable in connection with the present invention.

The voltage sample for the high order comparator is derived from the output of amplifier 41 at the junction of a pair of

capacitors

48 and 49 connected in series across the amplifier output. The current sample is taken from a coupling coil 50 which illustratively may be a ferrite ring surrounding the wire feeding RF energy to the pi network 37. The ferrite ring has a secondary winding 50a across which is connected a resistor 51. The current sample then is the voltage appearing across the resistor 51 which is proportional to the current flowing into the filter network 37. Other suitable means for sensing current may be used and will readily suggest themselves to those skilled in the art.

The voltage sample corresponds to the signal V described earlier in connection with FIG. 3. The sum of the current sample and voltage sample corresponds to signal V described earlier. These signals are supplied to a comparator corresponding to H0. 3.

The voltage sample, V,,, is impressed across a series circuit formed by a pair of parallel, relatively reversed

diodes

59, 60, a capacitor 63, a bypassed half of the center-tapped primary of an output transformer 69 and the secondary of transformer 65. As in FIG. 3, the input voltage, V is in series with a reference voltage V supplied from transformer 65 which is coupled to the output of a reference oscillator 73.

Similarly, the sum of the voltage and current samples, V is impressed across a series circuit formed by another pair of parallel, relatively reversed

diodes

61, 62, a capacitor 66, the other similarly bypassed half of the center-tapped primary of output transformer 69, and the transformer 65 secondary. This places the other input voltages V,, also in series with the reference voltage V as in FIG. 3.

The reference frequency is selected preferably in the audiofrequency range (illustratively kilohertz), while the RF samples are in the megahertz range as is conventional. This again proves to be of definite advantage in that the output signal appearing at the transformer 69 secondary is in the audiofrequency range, making it simple by the use of bypassed

capacitors

67, 68, to greatly attenuate modulation products at radiofrequency. Note how this differs markedly from the typical modulator whose output is a carrier plus certain sidebands.

In operation, the two RF signals related to the voltage and current applied to the tunable filter 37 are fed to the high order comparator. As stated previously, a sample of the RF voltage V is supplied to one input of the comparator and the sum of the RF voltage and current samples, is applied to the other input. The voltage and current samples are designed or adjusted (by any conventional means) so that V,, equals V when the system is at null. This can readily be done by substituting a test resistor for the network while making voltages V,, and V equal. One way of achieving this condition is to establish a selected voltage sample level for

V using capacitors

48 and 49 and then selecting resistor 51 to provide the desired voltageproportional to the current sample.

As is well known, when a filter is made to change its tuning condition, there is a relative phase shift between voltage and current which results in a marked change in amplitude of one relative to the other. This change is reflected in the inputs V and V This change is used in the comparator circuit to obtain a control signal for readjusting the filter to the desired condition of resonance.

As described above, the output of the comparator circuit will be a signal at the reference frequency W which will have null amplitude when voltages V and V are of equal amplitude. That output signal will reverse in polarity (i.e., phase sense) according as V, exceeds or is less than V and will vary in amplitude in correspondence with the magnitude difference between V, and V That output (bypassed for higher harmonics by capacitor 70) is supplied to an audio amplifier 71 of any conventional form, which serves further to suppress any RF components. The amplified output of audio amplifier 71 is supplied as an input signal to a phase detector 72, which may be a synchronous detector of any conventional form. Detector 72 is also supplied with a local signal from reference oscillator 73, phase shifted 90 with respect to the reference of V supplied to the comparator circuit.

When the RF signals are in proper relationship, that is, when the amplitude of the vectorial sum of the current and voltage samples V equals the amplitude of the voltage sample V alone, the output of phase detector 72 is at a null condition. When the two signals V and V are unequal, the relative magnitudes of the voltage and current samples will differ producing an error signal of the same frequency as the carrier at the input of detector 72. Detector 72 then converts this alternating error signal into a positive or negative DC error signal ofa polarity depending on whether the voltage sample V was greater or less than V,,, and of an amplitude corresponding to the difference in amplitude between V, and V This error voltage is then employed by the tuning control system 74 in a conventional way to adjust the value of appropriate elements (L or C) of filter 37 to bring the pi network into a state leading to a tuned and matched condition. Illustrative means for accomplishing the tuning of filter 37 have been described in the previously mentioned patent to Beitman U.S. Pat. No. 3,390,337.

The circuit contains almost no critical components. The resistor 51 is not critical. The main criterion is to have the reactance of coil 50 much greater than the resistance of resistor 51. The size of

capacitors

48 and 49 will depend on the voltage level associated with the amplifier stage. For many tetrode amplifiers, the B+ voltage supply may exceed 1,000 volts and the peak voltage at the high side of capacitor 48 is almost double the B+ voltage. In that case, capacitor 49 is approximately three orders of magnitude larger than capacitor 48 to achieve a voltage sample at the junction of

capacitors

48 and 49 on the order of 1 volt. If the amplifier stage incorporated a transistor as the active element, the peak voltage at the high side of capacitor 48 is desirably much lower and the size ratio of the capacitors may be in the order of fifty to one. As can be readily understood by those skilled in the art, a circuit using comparator 33 may be similarly configured to establish phase information necessary to provide the final tuned and matched condition required for filter 37. When the filter is both tuned and matched a condition of maximum power transfer is achieved and the RF voltage will be in phase with the applied RF current. Such a circuit arrangement has many diverse applications which need not be further discussed here.

As will be obvious to those skilled in the art, the comparator described above has many applications. The comparison of two AC signals of the same or different frequencies and of very low signal levels can be made with high accuracy. In fact most applications warranting conversion of AC signals to DC signals and subsequent comparison of the DC signals can be effected by the high-order comparator of the present invention. Such a circuit may be required in diverse control systems and high-frequency communications systems among other uses.

The use of one high-order comparator to compare signal amplitude information and another to compare phase information can provide information with regard to impedance parameters which may be applicable to innumerable tuning and impedance adjusting or impedance measuring circuits.

While there has been described what are at present considered to be the preferred embodiments of the invention, it will be obvious to those skilled in the art that various changes and modifications may be made therein and it is therefore intended to cover all such changes and modifications as fall within the true spirit and scope of the invention.

What is claimed is:

l. A mixer circuit for transforming amplitude variations of a first signal of a predetermined first frequency into corresponding amplitude variations of a second signal of a different frequency comprising:

nonlinear means for producing an output signal which is an odd, substantially exponential function of its input signal; means supplying only the AC portion of said first signal to said nonlinear means as an input signal thereto;

means also supplying a second signal of a frequency different from said first frequency to said nonlinear means as an input signal thereto; and

means for deriving an output from only those components of said nonlinear means output signal which are substantially of a multiple including unity of said second signal frequency.

2. A' circuit as described in claim 1 including means for supplying the sum of the AC portions of said first and second signals to said nonlinear means as an input signal.

3. A circuit as described in claim 1 wherein the frequency of said second signal is substantially lower than said first signal frequency.

4. A circuit as described in claim 1 wherein said nonlinear means comprises a pair of parallel-connected, oppositely polarized diodes whose total current is substantially an odd, exponential function of the voltage applied across said diodes.

5. A circuit as described in claim 1 wherein said multiple of said second signal frequency is approximately proportional to 1 plus 400 times the square of the amplitude of said first signal frequency.

6. A circuit as described in claim 1 wherein said means supplying only the AC portion of said first signal to said nonlinear means comprises a blocking capacitor.

7. A comparison circuit for producing a variable-amplitude alternating output signal corresponding to the difference between the amplitudes of two variable-amplitude alternating input signals comprising:

first and second nonlinear means, each for producing an output signal which is an odd, substantially exponential function of its input signal;

means for supplying to said first nonlinear means an input signal which substantially corresponds to the sum of an alternating reference signal and one of said variable-amplitude alternating input signals, to produce a first output signal;

means for supplying to said second nonlinear means an input signal which substantially corresponds to the sum of said reference signal and the other of said variable-amplitude alternating input signals to produce a second output signal; and

means for deriving a resultant output signal corresponding to the algebraic difference between said variable-amplitude alternating input signals.

8. A circuit as described in claim 7 wherein said first and second nonlinear means each comprise a pair of parallel-connected, oppositely polarized diodes whose total current is substantially an odd, exponential function of the voltage applied across said diodes.

9. A circuit as described in claim 7 wherein said supplying means each include a series capacitor for blocking DC signals.

10. A circuit as described in claim 7 wherein said deriving means includes means for suppressing signal components in said resultant output signal substantially of the frequency of each of said two variable-amplitude input signals.

ll. A circuit as described in claim 7 wherein said two variable-amplitude alternating input signals are at a substantially higher frequency than said variable-amplitude alternating out put signal.

12. A circuit as described in claim 7 wherein said means for deriving an output signal includes a transformer having a primary winding connected in series at one end to said first nonlinear means and connected in series at the other end to said second nonlinear means.

13. A circuit as described in claim 12 wherein said primary winding has a center-tap and wherein said reference signal is supplied to said center-tap.

14. A circuit as described in claim 7 wherein said resultant output signal is substantially of a multiple including unity of the reference signal.

l5. ln an automatic tuning system having a tunable circuit and it first signal corresponding to the voltage applied to said tunable circuit and a second signal related to the current flow to said tunable circuit, an amplitude comparison circuit llll adapted to have said first and second signals and a reference signal of a different frequency supplied thereto comprising:

first and second nonlinear means, each for producing an output which is an odd, substantially exponential function of input signal; means for supplying to said first nonlinear means an input signal which substantially corresponds to the sum of said first signal and said reference signal to produce a first out put signal;

means for supplying to said second nonlinear means an input signal which substantially corresponds to the sum of said second signal and said reference: signal to produce a second output signal; and

means for deriving a resultant output signal corresponding to the algebraic difference between said first and second signals;

whereby said resulting output signal may be used to tune said tunable circuit.

16. A circuit as described in claim 15 whereby said first and second nonlinear means each comprises a pair of parallel-connected, oppositely polarized diodes whose total current is substantially an odd, exponential function of the voltage applied across said diodes. I

17. A circuit as described in claim 15 wherein said means for deriving a resultant output signal includes an output transformer and a detector for producing a DC error signal.

18. A circuit as described in claim 15 wherein said supplying means each include a series capacitor for blocking DC signals.

19. In an automatic tuning system having a tunable circuit and a first signal corresponding to the voltage applied to said tunable circuit and a second signal related to the current flow to said tunable circuit, amplitude comparison circuit adapted to have said first and second signals and a reference signal of a different frequency supplied thereto comprising:

a first and second series capacitor each responsive to said first and second signals respectively for blocking DC signals;

a first pair of oppositely polarized, parallel-connected diodes connected in series with said first capacitor and responsive to the AC portion of said first input signal and to said reference signal for providing a first output signal which is an odd, substantially exponential function of the voltage developed across said first pair of diodes;

a second pair of oppositely polarized, parallel-connected diodes connected in series with the second blocking capacitor and responsive to the AC portion of said second input signal and to said reference signal for providing a second output signal which is an odd, substantially exponential function of the voltage developed across said second pair of diodes;

a centertapped output transformer, having the series circuit formed by said first blocking capacitor and first diode pair connected to one outer terminal and the series circuit formed by said second blocking capacitor and second diode pair connected to the other outer terminal and having the reference signal applied to said center-tapped ter minal, said transformer being responsive to said first and second current for developing an output voltage of the frequency of the reference signal with an amplitude corresponding to the algebraic difference between the squares of the individual first and second signals;

a pair of capacitors each connected across one half of the primary of the output transformer for filtering undesired signals and means responsive to said output voltage for detecting said voltage using said reference signal to produce a DC error signal whereby said l)( signal may be used to tune said tunable circuit.

Claims

1. A mixer circuit for transforming amplitude variations of a first signal of a predetermined first frequency into corresponding amplitude variations of a second signal of a different frequency comprising: nonlinear means for producing an output signal which is an odd, substantially exponential function of its input signal; means supplying only the AC portion of said first signal to said nonlinear means as an input signal thereto; means also supplying a second signal of a frequency different from said first frequency to said nonlinear means as an input signal thereto; and means for deriving an output from only those components of said nonlinear means output signal which are substantially of a multiple including unity of said second signal frequency.

2. A circuit as described in claim 1 including means for supplying the sum of the AC portions of said first and second signals to said nonlinear means as an input signal.

7. A comparison circuit for producing a variable-amplitude alternating output signal corresponding to the difference between the amplitudes of two variable-amplitude alternating input signals comprising: first and second nonlinear means, each for producing an output signal which is an odd, substantially exponential function of its input signal; means for supplying to said first nonlinear means an input signal which substantially corresponds to the sum of an alternating reference signal and one of said variable-amplitude alternating input signals, to produce a first output signal; means for supplying to said second nonlinear means an Input signal which substantially corresponds to the sum of said reference signal and the other of said variable-amplitude alternating input signals to produce a second output signal; and means for deriving a resultant output signal corresponding to the algebraic difference between said variable-amplitude alternating input signals.

11. A circuit as described in claim 7 wherein said two variable-amplitude alternating input signals are at a substantially higher frequency than said variable-amplitude alternating output signal.

15. In an automatic tuning system having a tunable circuit and a first signal corresponding to the voltage applied to said tunable circuit and a second signal related to the current flow to said tunable circuit, an amplitude comparison circuit adapted to have said first and second signals and a reference signal of a different frequency supplied thereto comprising: first and second nonlinear means, each for producing an output which is an odd, substantially exponential function of input signal; means for supplying to said first nonlinear means an input signal which substantially corresponds to the sum of said first signal and said reference signal to produce a first output signal; means for supplying to said second nonlinear means an input signal which substantially corresponds to the sum of said second signal and said reference signal to produce a second output signal; and means for deriving a resultant output signal corresponding to the algebraic difference between said first and second signals; whereby said resulting output signal may be used to tune said tunable circuit.

16. A circuit as described in claim 15 whereby said first and second nonlinear means each comprises a pair of parallel-connected, oppositely polarized diodes whose total current is substantially an odd, exponential function of the voltage applied across said diodes.

19. In an automatic tuning system having a tunable circuit and a first signal corresponding to the voltage applied to said tunable circuit and a second signal related to the current flow to said tunable circuit, amplitude comparison circuit adapted to have said first and second signals and a reference signal of a different frequency supplied thereto comprising: a first and second series capacitor each responsive to said first and second signals respectively for blocking DC signals; a first pair of oppositely polarized, parallel-connected diodes connected in series with said firSt capacitor and responsive to the AC portion of said first input signal and to said reference signal for providing a first output signal which is an odd, substantially exponential function of the voltage developed across said first pair of diodes; a second pair of oppositely polarized, parallel-connected diodes connected in series with the second blocking capacitor and responsive to the AC portion of said second input signal and to said reference signal for providing a second output signal which is an odd, substantially exponential function of the voltage developed across said second pair of diodes; a center-tapped output transformer, having the series circuit formed by said first blocking capacitor and first diode pair connected to one outer terminal and the series circuit formed by said second blocking capacitor and second diode pair connected to the other outer terminal and having the reference signal applied to said center-tapped terminal, said transformer being responsive to said first and second current for developing an output voltage of the frequency of the reference signal with an amplitude corresponding to the algebraic difference between the squares of the individual first and second signals; a pair of capacitors each connected across one half of the primary of the output transformer for filtering undesired signals and means responsive to said output voltage for detecting said voltage using said reference signal to produce a DC error signal whereby said DC signal may be used to tune said tunable circuit.