|Veröffentlichungsdatum||10. Dez. 2013|
|Eingetragen||13. Juli 2011|
|Prioritätsdatum||13. Juli 2011|
|Auch veröffentlicht unter||DE112012002932B4, DE112012002932T5, US20130015843, WO2013015956A1|
|Veröffentlichungsnummer||13181926, 181926, US 8604777 B2, US 8604777B2, US-B2-8604777, US8604777 B2, US8604777B2|
|Erfinder||Michael C. Doogue, Shaun D. Milano|
|Ursprünglich Bevollmächtigter||Allegro Microsystems, Llc|
|Zitat exportieren||BiBTeX, EndNote, RefMan|
|Patentzitate (68), Nichtpatentzitate (72), Referenziert von (5), Klassifizierungen (14), Juristische Ereignisse (4)|
|Externe Links: USPTO, USPTO-Zuordnung, Espacenet|
This invention relates generally to current sensors with integrated current conductors and, more particularly, to such a current sensor for use in a current divider configuration.
As is known in the art, one type of conventional current sensor uses a magnetic field transducer (for example a Hall effect or magnetoresistive transducer) in proximity to a current conductor. The magnetic field transducer generates an output signal having a magnitude proportional to the magnetic field induced by a current that flows through the current conductor. Typically, the current sensor also includes circuitry to amplify and condition the output signal of the magnetic field transducer.
The magnetic field transducer and circuitry are sometimes provided as an integrated circuit (IC) in an IC package that also contains a current conductor. Illustrative current sensors of this type are sold under part numbers ACS712 and ACS758xCB by Allegro MicroSystems, Inc. of Worcester, Mass. 01615, the Assignee of the subject application.
Various parameters characterize the performance of current sensors, including sensitivity. Sensitivity is related to the magnitude of a change in output voltage from the Hall effect transducer in response to a sensed current. The sensitivity of a current sensor is related to a variety of factors. One important factor is the physical separation between the Hall effect element and the current conductor.
Integration of the current conductor into an IC package allows for close and precise positioning of the current conductor relative to the magnetic field transducer. However, the amount of current that can be routed through the current conductor is limited by the physical and thermal limitations of the IC package.
One technique for measuring current levels that exceed the current carrying capability of the current sensor is to physically split the current path between the integrated current conductor and an external shunt current conductor that is coupled in parallel with the integrated current conductor. This technique can also be used to increase the current transient survival capability of the current sensor (also referred to as the current sensor's “over-current capability”) by diverting some current away from the current sensor. With such a current divider arrangement, only a portion of the total current to be measured is carried by the integrated current conductor and the remainder of the current is carried by the external shunt current conductor. The external shunt current conductor can be implemented with a trace or layer on a printed circuit board (PCB) on which the current sensor is mounted or with a bus bar. Current conductors that are configured as current dividers are designed to achieve a known division of current so that measurement of the current carried by the integrated current conductor can be used to determine the total current.
While this type of arrangement can be used to increase current measurement levels of an application as well as current carrying capability of the current sensor, there are disadvantages. Namely, since less current will flow through the integrated current conductor, the resulting magnetic field signal level (i.e., the resolution) will be lower. Also, manufacturing and assembly tolerances, both at the device and at the board level, result in some variability in the division of current between the integrated current conductor and the shunt current conductor from the designed division of current. For example, the resistance of the current sensor package's lead frame can vary over the course of time due to production tolerances. The manufacturing process of soldering the current sensor package leads to a PCB trace is also very important, as higher solder resistance can cause more current to pass through the shunt conductor and less through the integrated current conductor of the current sensor. The thickness and width of PCB traces can also vary as a function of manufacturing tolerances. Such variability is undesirable since the total current cannot be accurately determined from the measurement of current carried by the integrated current conductor. In applications where accuracy requirements make it necessary to compensate for these variations, some current sensors allow the sensitivity to be programmed after assembly as is described in an Application Note AN295036, Rev. 3, of Allegro MicroSystems, Inc. entitled “Using Allegro Current Sensor ICs in Current Divider Configurations for Extended Measurement Range” (and published at http://www.allegromicro.com/en/Products/Design/an/an295036.pdf), which Application Note is incorporated by reference herein in its entirety.
In general, in one aspect, the invention is directed to an integrated circuit current sensor that allows for self-calibration in a current divider configuration. The current sensor includes an integrated current conductor, a magnetic field transducer, a controllable gain stage and a calibration controller. The integrated current conductor is adapted to receive a portion of a calibration current. The calibration current corresponds to a full scale current. The magnetic field transducer, responsive to the calibration current portion, provides a magnetic field signal having a magnitude proportional to a magnetic field generated by the calibration current portion. The controllable gain stage is configured to amplify the magnetic field signal with an adjustable gain to provide an amplified magnetic field signal. The calibration controller is responsive to a calibration command signal to adjust the adjustable gain of the controllable gain stage to a calibrated gain in order to provide the amplified magnetic field signal at a predetermined voltage level that corresponds to a desired current sensor output signal voltage level if the full scale current were received by the integrated current conductor.
Embodiments of the invention may include one or more of the following features. The calibration controller can include a comparator having a first input responsive to a full scale reference voltage indicative of the predetermined voltage level, a second input responsive to the amplified magnetic field signal, and an output at which a comparator output signal is provided in a first state when the amplified magnetic field signal is less than the full scale reference voltage and in a second state when the amplified magnetic field signal is greater than the full scale reference voltage. The calibration controller can further include a counter that is responsive to the calibration command signal to start counting, responsive to the comparator output signal to stop counting, and that provides a counter output signal to the controllable gain stage for adjusting the adjustable gain. The full scale reference voltage can be determined by the calibration command signal. The controllable gain stage can include a variable resistor controlled by the counter output signal from the calibration controller. The controllable gain stage can further include an adjustable reference voltage and the calibration controller can be responsive to a calibration command signal for adjusting the reference voltage to provide the amplified magnetic field signal at a predetermined voltage level corresponding to the calibration current when the calibration current is zero amps.
The foregoing features of the invention, as well as the invention itself, may be more fully understood from the following detailed description of the drawings, in which:
Other package styles that can accommodate some type of internal current conductor along with the die 20 may be used. Another possible package option will be described later with reference to
In the current divider configuration, a total current to be measured, labeled “ITot”, is applied to the current conductor 14. A current sense portion of the current, labeled “I1”, flows through the current sense conductor portion 14 a into the leads 18 a, 18 b, which are shown to be electrically coupled in parallel, through the loop portion (not shown), and out of leads 18 c, 18 d, which are also electrically coupled in parallel. The remainder of the total current, or shunt current labeled “I2”, flows through the shunt conductor portion 14 b, thus bypassing the current sensor 12. The path of the current is indicated by reference numeral 26, and includes a total current path 28 for total current ITot, and separate subpaths including a shunt path 30 for shunt current I2 and a current sense path 32 a, 32 b for current sense current I1.
In this configuration the current to be measured by the current sensor, I1, travels in and out of the current sensor 12 on the primary side. Inside the IC package, the IC die 20 is placed over but does not make contact with the integrated current conductor loop, thereby providing galvanic (voltage) isolation. Although different package technologies can be used, the use of a flip-chip assembly allows the magnetic field transducer 22 of the IC die 20 to be positioned in very close proximity to the internal current conductor loop so that the magnetic signal coupling is maximized.
As noted above in the Background, the disadvantages of using a current divider configuration include loss of current sensor resolution and variability in current division (i.e., mismatch between I1 and I2). Various prior techniques used to address these sources of error include system-level calibration (or compensation) and PCB trace trimming. A system-level calibration involves applying current to a current conductor designed for a desired current division, recording the current sensor's output and scaling that output in an external controller to the desired value. This method requires a more complex control system to be implemented by the user. Alternatively, the PCB trace can be laser trimmed to achieve a desired division of current. The PCB trim method is even more complex than system level calibration and would be time consuming in the application test environment, resulting in a high cost system. As yet another alternative, some current sensors allow device sensitivity to be programmed after device assembly (as mentioned in the above-referenced application note).
In contrast, according to the present invention, the current sensor 12 employs a self-calibration technique. The self-calibration feature allows the current sensor to calibrate itself, more specifically, to adjust sensitivity until the current sensor accurately reflects at its output a value indicative of the total amount current to be measured by an application having a given current divider configuration, that is, ITot. Sensitivity refers to the change in a current sensor's output in response to a change in current being sensed by the device. The sensitivity of the device is the product of the magnetic circuit sensitivity (in G/A) and the current sensor's linear amplifier gain (in mV/G). The sensitivity of the current sensor may be optimized by adjusting the gain. The term “full scale current”, as it is used herein, refers to the maximum level of current that would be sensed by the current sensor in absence of a shunt path, that is, the total current ITot. The term “full scale voltage” refers to the output voltage that corresponds to the full scale current. During the self-calibration, as will be described in fuller detail later, the current sensor 12 adjusts gain so that it provides an output signal that is the equivalent of the output signal that would be provided by the current sensor if a shunt path were not utilized, that is, if the current sensor were to sense the full scale current ITot. In this manner, the self-calibration allows for increased current transient survival, i.e., improved over-current capability, on the part of the current sensor (as achieved through the use of a current divider configuration), but unlike prior calibration approaches requires minimal input from the user as well as provides a simpler and less expensive solution to the problems of a shunt system design.
One or more additional control lines, e.g., line 46 (shown in dashed lines) may be provided for calibration purposes as well. For example, the controller 34 may be configured to use control line 36 to initiate self-calibration, either through the type of signaling described above, or by using a simple serial calibration command, and to use a separate control line, line 46, to set values of certain calibration-required reference voltages to user-selected values, as will be described in further detail later. All of the calibration control could be provided to the current sensor using the same control signal line as well. A single control line like line 36 may be used to control/initiate different “phases” of self-calibration, or separate lines like lines 36 and 46 may be used as dedicated control lines to control/initiate a particular phase, as will be discussed later with reference to
The circuitry 56 operates to receive an output of the sensing element 54, in this example a Hall voltage, and generate from it a current sensor output 58. For purposes of illustration, the circuitry 56 is partitioned into various functional blocks or stages, including a sensing interface (or magnetic field signal generating) stage 60, a calibration control stage 62 and an output stage 64.
The sensing interface stage 60 can be implemented to include a number of different components to amplify and condition the sensing element's magnetic field signal output. The illustrated architecture includes a dynamic offset cancellation circuit 66, an amplifier 68, trim circuits 70 and 72, and a filter 74 shown as a low pass filter. The dynamic offset cancellation circuit 66, which is coupled to the SE 54 by connections 76, provides a DC offset adjustment for DC voltage errors associated with the magnetic field signal produced by the sensing element 54. The dynamic offset cancellation circuit 66 is coupled to the amplifier 68, which amplifies an offset adjusted SE output signal 78 provided by the dynamic offset cancellation circuit 66. Accuracy is optimized through trimming of sensitivity and temperature response, via the sensitivity trim circuit 70 and sensitivity temperature coefficient trim circuit 72, respectively. The sensitivity trim circuit 70 permits adjustment of the gain of the amplifier 68. The sensitivity temperature coefficient trim circuit 72 permits adjustment of the gain of the amplifier 68 in order to compensate for gain variations due to temperature.
The output of the amplifier 68, or amplifier output 80, is coupled to the filter 74. The filter 74 can be a low pass filter, as shown, and/or a notch filter. The filter 74 is selected in accordance with a variety of factors, including but not limited to, a desired response time and a frequency spectrum of noise associated with the sensing element, the dynamic offset cancellation circuit and the amplifier. The filter produces at its output a filtered, magnetic field output signal 82, which is provided as an input to the calibration control stage 62. Other implementations of the sensing interface stage 60 are possible.
Still referring to
The output stage 64 is implemented in the illustrated architecture to provide an analog buffered output. The output stage 64, as depicted, includes a buffer amplifier 96 having a first (non-inverting) input 98 and a second input 100. An RC filter 102 is connected between the first input and the Vsig output 88 from the controllable gain stage 86. Applied to the second input 100 is a reference voltage developed by a resistive voltage divider 104 coupled between Vout 58 and ground.
The magnetic field sensor 53 can be any type of magnetic field sensor and is therefore not limited to Hall effect technology. Thus, the sensing element 54 may take a form other than that of a Hall effect element, such as a magnetoresistance (MR) element. The magnetic field sensor 53 is provided in the form of an IC or die containing a substrate on which the various circuit elements (including the sensing element 54) are formed. Although only one sensing element is shown, that sensing element 54 could be replaced by a pair of sensing elements connected in a differential arrangement, or multiple MR elements connected in a bridge circuit. Components to be included in the stage 60 can vary with the type of sensing technology that is chosen.
The current sensor, in a current divider (or shunt) configuration, as shown in
The current sensor 50 will have least one terminal (or pin or lead) to correspond to I1in and I1out, VCC (to connect to an external power supply), GND (to connect to ground), an input and output. For example, and referring back to the SOIC 12 shown in
Details of the calibration control stage 62, in particular, the controllable gain stage 86 and the calibration controller 84, according to an exemplary embodiment, are shown in
Still referring to
The calibration control signal or command is provided to the calibration controller (via the Vcal line 92) to initiate the self-calibration. Before the self-calibration is initiated, the user causes the desired, full scale calibration current ITot to be applied to the current conductor (current conductor 14 as shown in
Once the gain adjustment portion of the self-calibration has been completed, the interface 120 can permanently store the calibrated gain value 142 on the chip, e.g., in a nonvolatile memory such as an EEPROM 144, as shown, or in some alternative manner. The EEPROM 144, if used, will be coupled to the interface 122 and count 142 (of the counter 126) via lines 146 and 148, respectively. Once the adjusted gain has been saved, the self-calibration is complete. These interconnections 146 and 148 are used to accomplish a transfer of the calibrated gain value 142 between counter 126 and the nonvolatile memory.
During subsequent, non-calibrating operation of the current sensor in the same current divider configuration with ITot as the total current to flow through the current conductor and I1 the portion to be sensed by the current sensor, the current sensor uses the calibrated gain to provide at the output a signal that has been scaled from a voltage level corresponding to the sensed current to a predetermined voltage level corresponding to the total current ITot.
Certain implementation details of the calibration controller 84 and controllable gain stage 86 are a matter of design choice. For example, as shown in
It will be understood that the current sensor 50 could have more input pins so that the calibration command and specification of Vfs (via the adjustment to the voltage divider variable resistor 132) may be provided as separate control signals. The interface 120 can be implemented as a digital serial interface, such as Inter-Integrated Circuit (I2C), Single-Edge Nibble Transmission (SENT), Peripheral Sensor Interface 5 (PSI5) or Serial Peripheral Interface (SPI), or a simple RS232 interface. Thus, for example, the calibration command provided at input 92 and transfer of the count to nonvolatile memory (e.g., EEPROM 144) can be achieved with messages according to a selected one of these or other suitable protocols.
Thus, for a predetermined voltage level of Vfs and corresponding full scale current level of calibration current (both user-selected levels for a given current divider configuration), the current sensor 50 can self-calibrate to adjust an adjustable gain of the controllable gain stage 86 to scale the value of that stage's output, the amplified magnetic field signal Vsig 88 (and therefore, the output provided at Vout 58 (from FIG. 3)), to the predetermined voltage level of Vfs. For example, assuming that the magnetic circuit sensitivity is in the order of magnitude of approximately 10 Gauss per amp (which means that for every amp flowing through the sensor's integrated current conductor 52, 10 Gauss of field is generated and sensed by the Hall element 54) and that the desired output is 3V when 20 Amps is flowing through the current sensor, then the gain of the controllable gain output stage 62 is calculated to be (3V−1.65V)/20 A/10 G/A or 67.5 mV/A. If a total current, “ITot”, of 20 Amps is passed through the current divider system and the current is split perfectly at 10 A through the shunt path (“I2”) and 10 A through the sensor (“I1”) and the desired output voltage on Vout is 3V, then the gain would be adjusted by the self-calibration to be twice the value calculated above, or (3−1.65)V/10 A/10 G/A, or 135 mV/A.
Adjustable gain values for the 3V output example and different current level splitting are given below in Table 1. The perfect 10 A split between the shunt and current sensor is shown in row 11 of table.
Current splitting and required gain for a 3 V output
In practice, designing a current configuration system that would require that the self-calibration increase the gain to very high levels (e.g., gains above 300 mV/A) is not practical because the Hall transducer output generates a weak signal, which means lower resolution and signal to noise ratio (since both the signal and the noise are amplified as the gain increases). On the other hand, designing the system so that most of the current flows through the current sensor and not the shunt path would do little to improve the over-current capability of the current sensor (which, of course, is one of the advantages of using a shunt path). In the 20 Amp example above, the current sensor could amplify the signal with less than 5 A in the current sensor and greater than 15 A provided through the shunt path for a gain of approximately 300 mV/A as shown in Table 1. This would allow for a split of 5 A/20 A or a 25% to 75% split through the current sensor and shunt path, respectively, while compensating for any error in splitting the current as discussed above.
Normally, when no current is flowing through the current sensor, the value of Vref (at amplifier input 116,
Because QVO often changes as a function of the gain of the current sensor IC, it may be desirable to expand the self-calibration of the calibration control stage 62 to include a second calibration, a QVO calibration, to be performed following the gain adjustment calibration. Referring to
Once a self-calibration has been initiated, and a gain adjustment portion and calibrated gain store has been completed, the QVO calibration can begin. First the user must set the current ITot through the system to 0 A and then send a second calibration command to the interface 120′. Upon receipt of the calibration command, the interface signals (via line 194) to the counter 180 to start counting. The count of the counter 180 changes and with each count iteration the value of variable resistor 188 is adjusted (thereby adjusting the value of Vref that is provided to the amplifier 110) until the comparator 178 determines that voltage level or magnitude of Vsig has passed the threshold provided by VQVO. That is, the comparator output signal 196 is provided in a first state when the magnitude of the amplified magnetic field signal Vsig (provided at 186) is less than that of the VQVO signal 184 and in a second state when the magnitude of the amplified magnetic field signal Vsig is greater than the magnitude of the VQVO signal 184. Once the value of Vsig has reached VQVO, the voltage reference adjustment portion of this second phase of the self-calibration is complete and the “final” count value from the counter 180 is saved in the EEPROM 144 or other suitable nonvolatile memory (or, alternately, a fuse network like fuse network 150 shown in
The sequencing of these two phases of the self-calibration activity is managed externally through control signals or commands provided by an external controller. A single control line or separate control lines, one to control the gain adjustment and the other to control the offset adjustment, may be used. The self-calibration may be repeated if desired. Also, a QVO calibration could be performed before and after the gain adjustment to provide an initial setting for the adjustable reference voltage and an updated setting following the gain adjustment. The QVO calibration, if performed at least once (following a gain adjustment), would increase the accuracy of the current sensor, especially for designs in which the gain is increased to high levels, e.g., gains above a 300 mV/A level. The QVO calibration could be performed only prior to the gain calibration but with less accurate results (for the reasons discussed above).
To support higher current measurements, another current sensor package option having a thicker current conductor may be used. An example is provided in
In sum, the current sensor with integrated current conductor and self-calibration as described above can be controlled (with minimal external control) to calibrate itself to have a calibrated gain that results in the current sensor output signal having a voltage level that corresponds to the full scale calibration current, not the sensed portion of the calibration current. In this manner, the self-calibrating current sensor can, with relative ease, compensate for lower signal resolution and “calibrate out” any current mismatch associated with a current divider configuration.
All references cited herein are hereby incorporated herein by reference in their entirety.
Having described preferred embodiments, which serve to illustrate various concepts, structures and techniques, which are the subject of this patent, it will now become apparent to those of ordinary skill in the art that other embodiments incorporating these concepts, structures and techniques may be used. Accordingly, it is submitted that that scope of the patent should not be limited to the described embodiments but rather should be limited only by the spirit and scope of the following claims.
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|Zitiert von Patent||Eingetragen||Veröffentlichungsdatum||Antragsteller||Titel|
|US8947101 *||4. Jan. 2013||3. Febr. 2015||Linear Technology Corporation||Method and system for measuring the resistance of a resistive structure|
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|US-Klassifikation||324/202, 73/514.16, 324/117.00H, 361/93.6, 324/117.00R, 324/126, 323/277, 324/130, 73/514.31, 324/219|
|Unternehmensklassifikation||G01R15/148, G01R15/207, G01R35/005|
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