WO2000014887A1 - Circuit electronique de correction de distorsion de transmodulation - Google Patents

Circuit electronique de correction de distorsion de transmodulation Download PDF

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Publication number
WO2000014887A1
WO2000014887A1 PCT/US1999/020613 US9920613W WO0014887A1 WO 2000014887 A1 WO2000014887 A1 WO 2000014887A1 US 9920613 W US9920613 W US 9920613W WO 0014887 A1 WO0014887 A1 WO 0014887A1
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WO
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Prior art keywords
signal
circuit
distortion correction
correction circuit
path
Prior art date
Application number
PCT/US1999/020613
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English (en)
Inventor
Martin Regehr
Original Assignee
Ortel Corporation
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Ortel Corporation filed Critical Ortel Corporation
Priority to AU59135/99A priority Critical patent/AU5913599A/en
Publication of WO2000014887A1 publication Critical patent/WO2000014887A1/fr

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Classifications

    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/34—Negative-feedback-circuit arrangements with or without positive feedback
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/20—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers
    • H03F3/24—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers of transmitter output stages
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/02—Transmitters
    • H04B1/04—Circuits
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/02—Transmitters
    • H04B1/04—Circuits
    • H04B2001/0408—Circuits with power amplifiers
    • H04B2001/0433—Circuits with power amplifiers with linearisation using feedback

Definitions

  • the present invention relates to predistorters for electronic and optical signals, and more particularly, to an electronic circuit for correcting cross modulation distortion.
  • Cross modulation distortion can occur in systems where several independently modulated RF carrier signals are transmitted along a shared transmission path.
  • one of the several carriers is intentionally amplitude modulated, the other carriers become unintentionally amplitude or phase modulated due to non-linearities in the transmission path.
  • Gain compression is one mechanism by which elements in a transmission path can generate cross modulation distortion. Gain compression occurs because an amplifier transmitting an RF signal typically produces less gain at higher RF signal power than at lower RF signal power. The amount of gain compression that occurs depends on the total RF power being transmitted. In a case where the amplifier is transmitting two carriers, one unmodulated at the amplifier input and the other amplitude modulated, unintentional amplitude modulation may occur in the first carrier since the gain it experiences depends on the total RF power which includes the amplitude of the second carrier. Other mechanisms may also generate cross modulation distortion.
  • the phase delay through an amplifier may also depend on the total RF power being transmitted through it. Accordingly, unintentional phase modulation can also occur as a result of the amplitude modulation on another carrier.
  • Cross modulation is essentially a third order distortion phenomenon, like CTB (composite triple beat).
  • CTB consists of third order intermodulation products of the carriers (three at a time), and so too cross modulation can be viewed as intermodulation between the modulated carrier, a modulation sideband, and the unmodulated carrier. Since cross modulation is a third order process, most circuits which produce CTB also produce cross modulation distortion, and predistorters for reducing or eliminating CTB typically have the effect of also reducing cross modulation distortion.
  • an electronic circuit for correcting cross modulation distortion in an RF transmission system.
  • the circuit comprises a modulator and an RF power monitor coupled along an RF transmission path.
  • a signal conditioning circuit which may include an amplifier, is coupled on a path to the modulator and RF power monitor.
  • the RF power monitor detects fluctuations in the RF power and produces a signal that depends on the RF power.
  • the signal is conditioned and fed to the modulator, which modulates the RF signal at appropriate levels to cancel cross modulation distortion produced by other elements of the system.
  • FIG. 1 is a general block diagram of an exemplary electronic circuit according to the present invention for correcting cross modulation distortion
  • FIG. 2 is a circuit diagram of the presently preferred embodiment of an RF power monitor used in the circuit of FIG. 1 ;
  • FIG. 3 is a circuit model which illustrates the operation of the Schottky diode used in the RF power monitor of FIG. 2;
  • FIG. 4 is a circuit diagram of a presently preferred embodiment of a modulator used in the circuit of FIG. 1 ;
  • FIG. 5 is a circuit diagram of an alternate embodiment of the modulator for modulating primarily higher frequencies
  • FIG. 6 is a circuit diagram of yet another alternate embodiment of the modulator for phase modulating the RF signal for correcting amplitude-to-phase cross modulation distortion
  • FIG. 7 is a block diagram of the presently preferred embodiment of the signal conditioning circuit used in the circuit of FIG. 1.
  • an electronic circuit 5 for correcting cross modulation is coupled to an RF transmission path 12 that includes one or more non-linear devices, such as RF amplifier 26, that produce cross modulation distortion by gain compression or other mechanisms.
  • the circuit 5 includes a modulator 16 and an RF power monitor 18 coupled to the RF transmission path, and a signal conditioning circuit 20 coupled between the RF power monitor and modulator through a path 22.
  • the non-linear device 26 may be upstream or downstream from the correction circuit or may even be located between the RF power monitor and the modulator as illustrated in this drawing.
  • the modulator is preferably placed at a point in the RF transmission path where the RF signal level is low whereas the RF power monitor is preferably placed at a point in the RF transmission path where the RF signal path is high.
  • the RF power monitor detects fluctuations in the RF power and generates an RF modulation signal that depends on the RF power; the signal is then conditioned and fed back to the modulator with the appropriate levels to cancel cross modulation distortion.
  • the RF amplifier is illustrated in FIG. 1 merely as an example of a possible non-linear device in the transmission path. Other devices producing cross modulation, several devices collectively producing cross modulation, or other causes of cross modulation, may be compensated by the techniques described herein.
  • the RF power monitor 18 in the preferred embodiment comprises a resistive splitter 30 (consisting of resistors Rl, R2, and R3) for diverting a small amount of the RF signal from the main RF transmission path, a Schottky diode circuit 32 (consisting of capacitors CI , C2, and C3, inductor LI, Schottky detector diode Dl and resistor R6 connected to bias voltage V B1AS ) for detecting the RF power, and a lossy resistive network 34 (consisting of resistors R4 and R5), coupled between the splitter and the diode circuit, for providing further isolation between the main transmission path and the diode.
  • a resistive splitter 30 consisting of resistors Rl, R2, and R3
  • a Schottky diode circuit 32 consististing of capacitors CI , C2, and C3, inductor LI, Schottky detector diode Dl and resistor R6 connected to bias voltage V B1AS
  • a directional coupler may be used instead of the resistive splitter 30.
  • a directional coupler which attenuates the throughpath by the same amount as a given resistive splitter will generally provide a larger RF level to the detector diode and smaller coupling of detector intermodulation distortion to the forward propagating RF transmission path.
  • non-linear elements in the cross-modulation correction circuit such as diodes, can generate unwanted intermodulation distortion directly from the RF signals flowing through them.
  • One or more conventional predistorter circuits may be used to cancel unwanted intermodulation distortion.
  • the resistive network of the splitter/attenuator preferably provides a sufficiently large RF level to the detector that the RF power measurement is accurate, while also providing enough isolation so that any intermodulation distortion generated by the detector does not compromise signal quality after propagating back through the resistive network into the main RF transmission path.
  • the diode circuit 32 includes resistor R6 and inductor LI for driving a DC bias current through the diode Dl to set its operating point.
  • the bias current is selected to provide the maximum sensitivity to changes in RF power.
  • the RF power monitor 18 preferably has acceptable sensitivity over the frequency range where cross modulation is to be suppressed.
  • the operation of the monitor can be understood qualitatively by considering the three types of current flowing through the Schottky diode Dl .
  • the RF current flows through resistors Rl , R3, R5 and capacitor
  • the RF modulation signal is generated by the Schottky diode due to its non-linear I-V characteristic and flows through inductor LI to ground, and through capacitor C3 to set up a voltage, at the output 35 of the circuit (corresponding to the input 68 of signal conditioning circuit 20 (FIG. 7)), that depends upon the detected RF power.
  • the diode can be replaced with a Norton equivalent circuit model, consisting of a current source 40 and an equivalent source impedance 42, which is equal to the impedance of the diode (Z SCH0TTKY ) at the modulation frequency when biased at its operating point (typically of order 1 kohm).
  • resistor R6 and the modulation-frequency impedance of capacitor C2 should preferably both be sufficiently large (compared to impedance Z SCH0TTKY ) not to load the diode excessively.
  • Inductor LI should preferably be sufficiently large to prevent the RF from shunting to ground but sufficiently small (relative to Z SCH0TTKY plus the input impedance of the signal conditioning circuit) to allow the modulation current to flow freely.
  • Capacitor CI should preferably be large enough (i.e., its impedance small compared to Z SCH0TTKY ) to allow the RF current to flow to the diode.
  • the output capacitor C3 should largely pass current at the modulation frequency. If appropriate, more specific band pass filter networks may be used.
  • the modulator 16 (which may be a voltage controlled attenuator) produces amplitude modulation which ideally exactly cancels the undesirable amplitude or phase modulation (or cross modulation) produced by the non-linear device. For example, if the gain of the device decreases by 0.01 dB when the RF level increases by 1 dB, then at the same time the modulator should be driven by the signal conditioning circuit, in response to the signal produced by the RF power monitor, so that its attenuation decreases by 0.01 dB, thereby maintaining the RF output signal at compensated full strength.
  • the modulator 16 contains first and second RF current diverting paths 41, 42 connected to the main RF transmission path adjacent to resistors R7 and R8 respectively.
  • the first current diverting path 41 includes a capacitor C4 connected in series to a resistor R9 and a PIN diode D2 to ground.
  • a modulator control signal from the signal conditioning circuit 20 (FIG. 7) is input at node 44 through capacitor C7 and resistor R10 to the first path.
  • the second current diverting path 42 includes a capacitor C5 connected in series to a resistor Rl 1.
  • the resistor Rl 1 is also connected to a parallel arrangement of a PIN diode D3 and a capacitor C6 connected to ground.
  • a second modulator control signal from the signal conditioning circuit 20 (FIG. 7) is input at node 46 through capacitor C8 and resistor R12 to the second path.
  • the modulator circuit 16 can also be understood by considering the three types of current flowing in it.
  • DC bias current flows through resistor R13, resistor R10, and diode D2 to ground, and through resistor R14, resistor R12, and diode D3 to ground.
  • Modulator control current flows through capacitor C7, resistor R10, and diode D2 to ground and through capacitor C8, resistor R12, and diode D3 to ground.
  • the RF signal current flows from RF IN through resistor R7 and resistor R8 to RF OUT.
  • Capacitors C7 and C8 are preferably chosen large enough to allow the modulation current to flow through them (i. e, their impedance preferably should be small compared to RIO and R12 respectively).
  • Capacitors C4 and C5 are preferably chosen large enough to allow the RF current to flow through them (i.e., their RF impedance preferably should be small compared to R9 and Rl 1, respectively), but small enough to prevent the modulation current from bypassing the diodes (i.e., the modulation-frequency impedance of C4 + R9 should be large compared to Z PIN the maximum impedance of the PIN diode D2).
  • Resistors R13 and R14 are similarly chosen large enough to prevent modulation current from bypassing the PIN diodes.
  • R9 and Rl 1 are determined by several factors, including the need for adequate attenuation range, any restrictions on insertion loss, and the need to prevent intermodulation distortion in the PIN diodes from degrading the signal. Larger values of R9 or Rl 1 reduce the minimum insertion loss of the modulator, and reduce the amount of intermodulation distortion contributed to the signal by the PIN diodes, but correspond to a smaller range of attenuations.
  • the DC bias points of PIN diodes D2 and D3 should preferably be chosen to maximize the RF modulation produced by the modulator control signal.
  • the circuit of FIG. 4 contains first and second diversion paths for compensating for different amounts of cross modulation distortion at different carrier frequencies.
  • capacitor C6 is used to allow the higher- frequency RF carriers to bypass diode D3, so that the modulator control signal applied to diode D3 will modulate primarily the lower-frequency carriers.
  • the first current diversion path 41 provides correction of cross modulation distortion over all frequencies, while the second current diversion path 42 provides an additional means for adjusting the amount of correction at lower frequencies. Therefore, in some cases, it may desirable to produce different amounts of correction dependent on RF signal frequencies.
  • the following values are used: R7_ R8 _ 1.5 ohms
  • an alternate embodiment of a current diversion path for modulating primarily higher frequency carriers includes a single current diversion path 50 connected between impedance matching resistors R15 and R16 on the main RF transmission path 12.
  • the current diversion path contains a capacitor C9 connected to a resistor R17 and PIN diode D4 to ground. This circuit is substantially similar to the first current diversion path 41 in the modulator of FIG.
  • capacitor C9 is preferably chosen to be sufficiently small (e.g., 0.5 pF) that it isolates the PIN diode D4 from the main RF signal path at low frequencies, thereby modulating the lower frequency carriers to a smaller extent than the high frequency carriers.
  • This current diversion path can be used in combination with either or both of the current diversion paths of FIG. 4 to form an alternate embodiment of the modulator.
  • a number of tuned or band pass type current diversion paths may be combined for obtaining a desired modulation which is different in different frequency bands to better match the cross modulation for a non-linear device or plurality of devices in the main signal path.
  • FIG. 6 another alternate modulator circuit is disclosed for cancellation of amplitude-to-phase cross-modulation in which intentional amplitude modulation of one carrier appears as phase modulation on another carrier.
  • the circuit includes an electrical path 60 connected to the main RF transmission path between impedance matching resistors R18 and R19.
  • the electrical path includes a capacitor CIO connected to a resistor R20 and a varactor diode D5 to ground.
  • the circuit accepts a modulation signal from the signal conditioning circuit through resistor R21 and node 62.
  • the varactor can be viewed as a voltage controlled capacitor.
  • the modulation signal modulates the capacitance of the diode, which in turn modulates the phase of the RF carriers.
  • the signal conditioning circuit 20 amplifies and filters the RF modulation signal such that when this signal is applied to the modulator 16 (FIG. 1), it produces modulation which ideally exactly cancels the cross modulation produced in other parts of the system. In some cases, amplification may not be necessary.
  • the signal conditioning circuit is a linear circuit that includes a voltage controlled adjustable gain amplifier which comprises first and second cascaded op-amps 70, 72 and two analog multipliers 74, 76. The first op-amp receives input from the RF power monitor 18
  • the first multiplier 74 has first and second inputs 74a and 74b.
  • the first input 74a is connected to the output of the second op-amp 72, and the second input 74b is adapted for receiving a control voltage VI at node 78.
  • the second multiplier 76 also has first and second inputs, 76a and 76b, in which the first input 76a is connected to the output of the second op- amp 72, and the second input 76b is adapted for receiving a second control voltage V2 at node 79.
  • the signal conditioning circuit has two outputs coupled to nodes 44 and 46 for providing modulator control voltages to the modulator 16 (FIG. 4, preferred embodiment).
  • the signal conditioning circuit can be altered as necessary for attachment to the alternate modulator circuits of FIGS. 5 and 6.
  • reactive elements such as capacitors can be added to the signal conditioning circuit to affect the frequency response of the circuit.
  • the analog multipliers provide for voltage-controlled adjustment of the gain and polarity of the corrective cross modulation.
  • the signal conditioning circuit is designed to have adjustable gain and selectable polarity, so that the amplitude and polarity of cross modulation produced by the correction circuit can be manually adjusted, or automatically adjusted by a microprocessor (not shown), by varying control voltages VI and V2.
  • the frequency response of the signal conditioning circuit is preferably designed to provide the desired amplitude and phase of corrective cross modulation over the modulation frequency range of interest. Depending on the mechanisms responsible for cross modulation, low frequency modulation of one carrier may be more strongly transferred to other carriers than high frequency modulation. In this case, the signal conditioning circuit should preferably be designed to have more gain at low frequencies than at high frequencies.
  • the signal conditioning circuit should also preferably compensate for any frequency dependence in the RF power monitor or in the voltage controlled modulator.
  • the PIN diode modulator in the preferred embodiment when driven with a 10 kHz current, will modulate the RF attenuation perhaps 4 dB more than when driven by a 100 kHz current; moreover, it is believed that the phase of the modulation imposed on the RF signal lags the phase of the modulation current.
  • the frequency response of the signal conditioning circuit should be designed to incorporate a corresponding phase lead and an increase in gain between 10 kHz and 100 kHz.
  • capacitor CIO and resistor R18 form a passive filter which introduces a phase lead and reduces the gain of the signal conditioning circuit at low frequencies.
  • the capacitor CIO also serves to prevent the DC component of the voltage generated by the RF power monitor from being amplified by the signal conditioning circuit.
  • a frequency-dependent response may also be incorporated into the signal conditioning circuit by using active filters. For example, the high-frequency gain of the signal conditioning circuit may be reduced by connecting a capacitor in parallel with the resistor in the feedback path of one of the op-amp circuits.
  • the signal conditioning circuit may be either linear or non-linear. It will be understood by those skilled in the art that the above described embodiments are merely illustrative of circuits that may be used to implement the present invention. For example, the relative locations of the modulator 16 and RF power monitor 18 may be varied. Normally, the amplitude modulation produced by the circuit is small, and the effect at the monitor of changes to the signal level due to the modulator is small. Therefore, the circuit will function regardless of whether the monitor precedes or comes after the modulator in the RF transmission path, and regardless of whether other RF transmitting elements are interposed between the RF power monitor and modulator on the RF transmission path.
  • the monitor it is preferable, however, to locate the monitor at a point in the transmission path where the signals are large, so the detector diode Dl (FIG. 2) can be adequately isolated from the main signal path while still receiving a sufficiently strong signal. It is also preferable to locate the modulator at a point where the signal is relatively weak, so that it can be strongly coupled to the signal path without introducing excessive distortion. In the preferred embodiment shown in FIG. 1 , this is accomplished by the illustrated placement of the monitor and modulator relative to the RF amplifier.

Abstract

L'invention concerne l'utilisation d'un circuit électronique pour corriger une distorsion de transmodulation. Une distorsion de transmodulation peut se produire dans des systèmes dans lesquels plusieurs signaux de porteuse à haute fréquence à modulation indépendante sont transmis le long d'une trajectoire de transmission partagée. Lorsqu'une des multiples porteuses subit une modulation intentionnelle en amplitude, les autres porteuses subissent une modulation non intentionnelle en amplitude et/ou en phase en raison de non-linéarités dans le canal de transmission. Le circuit est utilisé dans un système de transmission à haute fréquence (5) et comprend un modulateur (16) et un dispositif de contrôle (18) de puissance haute fréquence couplés le long d'une trajectoire (14) de transmission à haute fréquence. Un circuit (72) de mise en forme de signaux, pouvant comprendre un amplificateur (20) est couplé dans une trajectoire au dispositif de contrôle (18) de puissance haute fréquence et au modulateur. Pendant le fonctionnement, le dispositif de contrôle (18) de puissance haute fréquence détecte des fluctuations dans la puissance haute fréquence et produit un signal dépendant de la puissance haute fréquence. Le signal est mis en forme et introduit dans le modulateur (16) qui module le signal haute fréquence aux niveaux appropriés pour annuler une distorsion de transmodulation produite par d'autres éléments du système.
PCT/US1999/020613 1998-09-09 1999-09-09 Circuit electronique de correction de distorsion de transmodulation WO2000014887A1 (fr)

Priority Applications (1)

Application Number Priority Date Filing Date Title
AU59135/99A AU5913599A (en) 1998-09-09 1999-09-09 Electronic circuit for correcting cross modulation distortion

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
US15063498A 1998-09-09 1998-09-09
US09/150,634 1998-09-09

Publications (1)

Publication Number Publication Date
WO2000014887A1 true WO2000014887A1 (fr) 2000-03-16

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PCT/US1999/020613 WO2000014887A1 (fr) 1998-09-09 1999-09-09 Circuit electronique de correction de distorsion de transmodulation

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WO (1) WO2000014887A1 (fr)

Cited By (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US8320773B2 (en) 2008-10-03 2012-11-27 Applied Optoelectronics, Inc. Reducing cross-modulation in multichannel modulated optical systems
US9191111B2 (en) 2008-10-03 2015-11-17 Applied Optoelectronics, Inc. Reducing cross-modulation in multichannel modulated optical systems
CN113659938A (zh) * 2021-08-24 2021-11-16 电子科技大学 一种模拟预失真器

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US3952251A (en) * 1974-07-15 1976-04-20 Kahn Leonard R Narrow bandwidth, compatible single sideband (CSSB) transmission system, and three tone generator used therein
US4618999A (en) * 1983-02-23 1986-10-21 U.S. Philips Corporation Polar loop transmitter
US4933986A (en) * 1989-08-25 1990-06-12 Motorola, Inc. Gain/phase compensation for linear amplifier feedback loop
US5335369A (en) * 1990-05-21 1994-08-02 Kabushiki Kaisha Toshiba Automatic power control circuit for use with a radio telecommunication apparatus
US5420536A (en) * 1993-03-16 1995-05-30 Victoria University Of Technology Linearized power amplifier
US5430416A (en) * 1994-02-23 1995-07-04 Motorola Power amplifier having nested amplitude modulation controller and phase modulation controller
US5553318A (en) * 1994-09-30 1996-09-03 Nec Corporation Transmitter having envelope feedback loop and automatic level control loop
US5613226A (en) * 1993-11-30 1997-03-18 Nec Corporation Linear transmitter for use in combination with radio communication systems
US5732333A (en) * 1996-02-14 1998-03-24 Glenayre Electronics, Inc. Linear transmitter using predistortion

Patent Citations (9)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US3952251A (en) * 1974-07-15 1976-04-20 Kahn Leonard R Narrow bandwidth, compatible single sideband (CSSB) transmission system, and three tone generator used therein
US4618999A (en) * 1983-02-23 1986-10-21 U.S. Philips Corporation Polar loop transmitter
US4933986A (en) * 1989-08-25 1990-06-12 Motorola, Inc. Gain/phase compensation for linear amplifier feedback loop
US5335369A (en) * 1990-05-21 1994-08-02 Kabushiki Kaisha Toshiba Automatic power control circuit for use with a radio telecommunication apparatus
US5420536A (en) * 1993-03-16 1995-05-30 Victoria University Of Technology Linearized power amplifier
US5613226A (en) * 1993-11-30 1997-03-18 Nec Corporation Linear transmitter for use in combination with radio communication systems
US5430416A (en) * 1994-02-23 1995-07-04 Motorola Power amplifier having nested amplitude modulation controller and phase modulation controller
US5553318A (en) * 1994-09-30 1996-09-03 Nec Corporation Transmitter having envelope feedback loop and automatic level control loop
US5732333A (en) * 1996-02-14 1998-03-24 Glenayre Electronics, Inc. Linear transmitter using predistortion

Cited By (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US8320773B2 (en) 2008-10-03 2012-11-27 Applied Optoelectronics, Inc. Reducing cross-modulation in multichannel modulated optical systems
US9191111B2 (en) 2008-10-03 2015-11-17 Applied Optoelectronics, Inc. Reducing cross-modulation in multichannel modulated optical systems
CN113659938A (zh) * 2021-08-24 2021-11-16 电子科技大学 一种模拟预失真器
CN113659938B (zh) * 2021-08-24 2023-05-12 电子科技大学 一种模拟预失真器

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